Download: LM4780 LM4780 Overture Audio Power Amplifier Series Stereo 60W, Mono 120W Audio Power Amplifier with Mute Literature Number: SNAS193A

LM4780 LM4780 Overture Audio Power Amplifier Series Stereo 60W, Mono 120W Audio Power Amplifier with Mute Literature Number: SNAS193A LM4780 Overture™ Audio January 22, 2010 Power Amplifier Series Stereo 60W, Mono 120W Audio Power Amplifier with Mute General Description Key Specifications The LM4780 is an stereo audio amplifier capable of typically delivering 60W per channel of continuous average output ■ Output Power/Channel power into an 8Ω load with less than 0.5% THD+N from 20Hz with 0.5% THD+N, 1kHz into 8Ω 60W (typ) - 20kHz. ■ THD+N at2x30W into 8Ω (20Hz - 20kHz) 0.03% (typ) The LM4780 i...
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LM4780

LM4780 Overture Audio Power Amplifier Series Stereo 60W, Mono 120W Audio Power Amplifier with Mute Literature Number: SNAS193A,

LM4780 Overture™ Audio January 22, 2010 Power Amplifier Series Stereo 60W, Mono 120W Audio Power Amplifier with Mute General Description Key Specifications

The LM4780 is an stereo audio amplifier capable of typically delivering 60W per channel of continuous average output ■ Output Power/Channel power into an 8Ω load with less than 0.5% THD+N from 20Hz with 0.5% THD+N, 1kHz into 8Ω 60W (typ) - 20kHz. ■ THD+N at2x30W into 8Ω (20Hz - 20kHz) 0.03% (typ) The LM4780 is fully protected utilizing National's Self Peak ■ THD+N at2x30W into 6Ω (20Hz - 20kHz) 0.05% (typ) Instantaneous Temperature (°Ke) (SPiKeTM) protection cir- ■ THD+N at2x30W into 4Ω (20Hz - 20kHz) 0.07% (typ) cuitry. SPiKe provides a dynamically optimized Safe Operat- ■ Mute Attenuation 110dB (typ) ing Area (SOA). SPiKe protection completely safeguards the PSRR 85dB (min) LM4780's outputs against over-voltage, under-voltage, over- ■ loads, shorts to the supplies or GND, thermal runaway and ■ Slew Rate 19V/µs (typ) instantaneous temperature peaks. The advanced protection features of the LM4780 places it in a class above discrete and Features hybrid amplifiers. ■ SPiKe Protection Each amplifier of the LM4780 has an independent smooth ■ Low external component count transition fade-in/out mute. ■ Quiet fade-in/out mute mode The LM4780 can easily be configured for bridge or parallel Wide supply range: 20V - 84V operation for 120W mono solutions. ■ ■ Signal-to-Noise Ratio ≥ 97dB (ref. to PO = 1W)

Applications

■ Audio amplifier for component stereo ■ Audio amplifier for compact stereo ■ Audio amplifier for self-powered speakers ■ Audio amplifier for high-end and HD TVs

Typical Application

FIGURE 1. Typical Audio Amplifier Application Circuit Overture® is a registered trademark of National Semiconductor. © 2010 National Semiconductor Corporation 200586 www.national.com

LM4780 Overture® Audio Power Amplifier Series Stereo 60W, Mono 120W Audio Power Amplifier with Mute

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Connection Diagrams

Plastic Package (Note 14) 200586d6 Top View Order Number LM4780TA See NS Package Number TA27A TO-220 Top Marking 200586a2 Top View U - Wafer Fab Code Z - Assemble Plant Code XY - Date Code TT - Die Run Traceability L4780TA - LM4780TA www.national.com 2,

Absolute Maximum Ratings (Note 1, Note Junction Temperature (TJMAX) (Note 9) 150°C

Soldering Information 2) TA Package (10 seconds) 260°C If Military/Aerospace specified devices are required, Storage Temperature -40°C to +150°C please contact the National Semiconductor Sales Office/ Thermal Resistance Distributors for availability and specifications.  θJA 30°C/W Supply Voltage |V+| + |V-|  θJC 0.8°C/W (No Signal) 94V Supply Voltage |V+| + |V-| Operating Ratings (Note 1, Note 2) (Input Signal) 84V Temperature Range Common Mode Input Voltage (V+ or V-) and ≤ TMIN ≤ TA ≤ TMAX −20°C ≤ TA ≤ +85°C|V+| + |V-| 80V Supply Voltage |V+| + |V-| Differential Input Voltage (Note 13) 60V 20V ≤ VTOTAL ≤ 84V Output Current Internally Limited Note: Operation is guaranteed up to 84V; however, distortion Power Dissipation (Note 3) 125W may be introduced from SPiKe protection circuitry if proper ESD Susceptability (Note 4) 3.0kV thermal considerations are not taken into account. Refer to the Thermal Considerations section for more information. ESD Susceptability (Note 5) 200V

Electrical Characteristics (Note 1, Note 2)

The following specifications apply for V+ = +35V, V- = −35V, IMUTE = -1mA and RL = 8Ω unless otherwise specified. Limits apply for TA = 25°C. LM4780 Typical Limit Units Symbol Parameter Conditions (Note 6) (Note 7, (Limits) Note 8) Power Supply Voltage ≥ 20 V (min)|V+| + |V-| GND − V- 9V 18 (Note 10) 84 V (max) AM Mute Attenuation IMUTE = 0mA 110 80 dB (min) THD+N = 0.5% (max) f = 1kHz; f = 20kHz P Output Power (RMS) |V+| = |V-| = 25V, RL = 4ΩO 55 50 W (min) |V+| = |V-| = 30V, RL = 6Ω 55 50 W (min) |V+| = |V-| = 35V, RL = 8Ω 60 50 W (min) PO = 30W, f = 20Hz - 20kHz AV = 26dB Total Harmonic Distortion + THD+N |V+| = |V-| = 25V, RL = 4ΩNoise 0.07 % |V+| = |V-| = 30V, RL = 6Ω 0.05 % |V+| = |V-| = 35V, RL = 8Ω 0.03 % PO = 10W, f = 1kHz 70 dB Xtalk Channel Separation (Note 11) PO = 10W, f = 10kHz 72 dB SR Slew Rate VIN = 2.0VP-P, tRISE = 2ns 19 8 V/μs (min) Total Quiescent Power VCM = 0V, 110 170 mA (max)IDD Supply Current VO = 0V, IO = 0A VOS Input Offset Voltage VCM = 0V, IO = 0mA 1 10 mV (max) IB Input Bias Current VCM = 0V, IO = 0mA 0.2 1 μA (max) IO Output Current Limit |V+| = |V-| = 20V, tON = 10ms 11.57A(min) Output Dropout Voltage |V+ - VO|, V+ = 28V, IO = +100mA 1.6 2.0 V (max)VOD (Note 12) |V- - VO|, V- = -28V, IO = -100mA 2.5 3.0 V (max) V+ = 40V to 20V, V- = -40V, 120 85 dB (min) Power Supply Rejection Ratio VCM = 0V, IO = 0mA

PSRR

(Note 15) V+ = 40V, V- = -40V to -20V, 105 85 dB (min) VCM = 0V, IO = 0mA 3 www.national.com,

LM4780 Typical Limit Units Symbol Parameter Conditions

(Note 6) (Note 7, (Limits)

Note 8) Common Mode Rejection Ratio V+ = 60V to 20V, V- = -20V to -60V, CMRR 110 85 dB (min)

(Note 15) VCM = 20V to -20V, IO = 0mA

AVOL Open Loop Voltage Gain RL = 2kΩ, ΔVO = 40V 115 90 dB (min) GBWP Gain Bandwidth Product fIN = 100kHz, VIN = 50mVRMS82MHz (min) IHF-A-Weighting Filter,

eIN Input Noise = 600Ω ( 2.0 10 μV (max)RIN Input Referred)

PO = 1WRMS; A-Weighted Filter

97 dB fIN = 1kHz, RS = 25Ω

SNR Signal-to-Noise Ratio PO = 50WRMS; A-Weighted Filter

114 dB fIN = 1kHz, RS = 25Ω 60Hz, 7kHz, 4:1 (SMPTE) 0.004 %

IMD Intermodulation Distortion

60Hz, 7kHz, 1:1 (SMPTE) 0.009 % Note 1: All voltages are measured with respect to the ground pins, unless otherwise specified. Note 2: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions which guarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters where no limit is given; however, the typical value is a good indication of device performance. Note 3: The maximum power dissipation must be de-rated at elevated temperatures and is dictated by TJMAX, θJC, and the ambient temperature TA. The maximum allowable power dissipation is PDMAX = (TJMAX -TA) / θJC or the number given in the Absolute Maximum Ratings, whichever is lower. For the LM4780, TJMAX = 150°C and the typical θJC is 0.8°C/W. Refer to the Thermal Considerations section for more information. Note 4: Human body model, 100pF discharged through a 1.5kΩ resistor. Note 5: Machine Model: a 220pF - 240pF discharged through all pins. Note 6: Typical specifications are measured at 25°C and represent the parametric norm. Note 7: Tested limits are guaranteed to National's AOQL (Average Outgoing Quality Level). Note 8: Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis. Note 9: The maximum operating junction temperature is 150°C. However, the instantaneous Safe Operating Area temperature is 250°C. Note 10: V- must have at least - 9V at its pin with reference to GND in order for the under-voltage protection circuitry to be disabled. In addition, the voltage differential between V+ and V- must be greater than 14V. Note 11: Cross talk performance was measured using the demo board shown in the datasheet. PCB layout will affect cross talk. It is recommended that input and output traces be separated by as much distance as possible. Return ground traces from outputs should be independent back to a single ground point and use as wide of traces as possible. Note 12: The output dropout voltage is defined as the supply voltage minus the clipping voltage. Refer to the Clipping Voltage vs. Supply Voltage graph in the Typical Performance Characteristics section. Note 13: The Differential Input Voltage Absolute Maximum Rating is based on supply voltages V+ = 40V and V- = - 40V. Note 14: The TA27A is a non-isolated package. The package's metal back, and any heat sink to which it is mounted are connected to the V- potential when using only thermal compound. If a mica washer is used in addition to thermal compound, θCS (case to sink) is increased, but the heat sink will be electrically isolated from V-. Note 15: DC electrical test. Note 16: CCIR/ARM: A Practical Noise Measurement Method; by Ray Dolby, David Robinson and Kenneth Gundry, AES Preprint No. 1353 (F-3). www.national.com 4,

Bridged Amplifier Application Circuit

FIGURE 2. Bridged Amplifier Application Circuit 5 www.national.com,

Parallel Amplifier Application Circuit

FIGURE 3. Parallel Amplifier Application Circuit www.national.com 6,

Single Supply Application Circuit

*Optional components dependent upon specific design requirements. FIGURE 4. Single Supply Amplifier Application Circuit

Auxiliary Amplifier Application Circuit

200586d5 FIGURE 5. Special Audio Amplifier Application Circuit 7 www.national.com,

External Components Description

(Figures 1-5) Components Functional Description Prevents current from entering the amplifier's non-inverting input. This current may pass through to the load 1 RB during system power down, because of the amplifier's low input impedance when the undervoltage circuitry is off. This phenomenon occurs when the V+ and V- supply voltages are below 1.5V. 2 Ri Inverting input resistance. Along with Rf, sets AC gain. 3 Rf Feedback resistance. Along with Ri, sets AC gain. Feedback resistance. Works with Cf and Rf creating a lowpass filter that lowers AC gain at high frequencies.

R

4 f2 The -3dB point of the pole occurs when: (Rf - Ri)/2 = Rf // [1/(2πfcCf) + Rf2] for the Non-Inverting configuration(Note 17) shown in Figure 5.

C

5 f Compensation capacitor. Works with Rf and Rf2 to reduce AC gain at higher frequencies.(Note 17) CC Compensation capacitor. Reduces the gain at higher frequencies to avoid quasi-saturation oscillations of the6 (Note 17) output transistor. Also suppresses external electromagnetic switching noise created from fluorescent lamps. C Feedback capacitor which ensures unity gain at DC. Along with Ri also creates a highpass filter at fc = 1/ 7 i (Note 17) (2πRiCi). Provides power supply filtering and bypassing. Refer to the Supply Bypassing application section for proper 8 CS placement and selection of bypass capacitors.

R

9 V Acts as a volume control by setting the input voltage level. (Note 17) Sets the amplifier's input terminals DC bias point when CIN is present in the circuit. Also works with CIN to create

R

10 IN a highpass filter at fC = 1/(2πRINCIN). If the value of RIN is too large, oscillations may be observed on the outputs(Note 17) when the inputs are floating. Recommended values are 10kΩ to 47kΩ. Refer to Figure 5.

C

11 IN Input capacitor. Prevents the input signal's DC offsets from being passed onto the amplifier's inputs. (Note 17)

R

12 SN Works with CSN to stabilize the output stage by creating a pole that reduces high frequency instabilities.(Note 17) C Works with RSN to stabilize the output stage by creating a pole that reduces high frequency instabilities. The 13 SN (Note 17) pole is set at fC = 1/(2πRSNCSN). Refer to Figure 5. 14 L (Note 17) Provides high impedance at high frequencies so that R may decouple a highly capacitive load and reduce the Q of the series resonant circuit. Also provides a low impedance at low frequencies to short out R and pass audio 15 R (Note 17) signals to the load. Refer to Figure 5. 16 RA Provides DC voltage biasing for the transistor Q1 in single supply operation. 17 CA Provides bias filtering for single supply operation. RINP Limits the voltage difference between the amplifier's inputs for single supply operation. Refer to the Clicks and18 (Note 17) Pops application section for a more detailed explanation of the function of RINP. Provides input bias current for single supply operation. Refer to the Clicks and Pops application section for a 19 RBI more detailed explanation of the function of RBI. Establishes a fixed DC current for the transistor Q1 in single supply operation. This resistor stabilizes the half- 20 RE supply point along with CA. Mute resistance set up to allow 0.5mA to be drawn from each MUTE pin to turn the muting function off. 21 RM → RM is calculated using: RM ≤ (|VEE| − 2.6V)/l where l ≥ 0.5mA. Refer to the Mute Attenuation vs Mute Current curves in the Typical Performance Characteristics section. 22 CM Mute capacitance set up to create a large time constant for turn-on and turn-off muting. 23 S1 Mute switch. When open or switched to GND, the amplifier will be in mute mode. 24 ROUT Reduces current flow between outputs that are caused by Gain or DC offset differences between the amplifiers. Note 17: Optional components dependent upon specific design requirements. www.national.com 8, is the coupling capacitor, CC, and the compensation capaci-

Optional External Component tor, Cf. These two components are low impedances at certain

frequencies. They may couple signals from the input to the

Interaction output. Please take careful note of basic amplifier component

The optional external components have specific desired func- functionality when selecting the value of these components tions. Their values are chosen to reduce the bandwidth and and their placement on a printed circuit board (PCB). eliminate unwanted high frequency oscillation. They may, The optional external components shown in Figure 4 and however, cause certain undesirable effects when they inter- Figure 5, and described above, are applicable in both single act. Interaction may occur when the components produce and split supply voltage configurations. reactions that are nearly equal to one another. One example

Typical Performance Characteristics

THD+N vs Frequency THD+N vs Frequency ± 25V, POUT = 1W & 30W/Channel ± 30V, POUT = 1W & 30W/Channel RL= 4Ω, 80kHz BW RL= 6Ω, 80kHz BW 200586e3 200586e4 THD+N vs Frequency THD+N vs Output Power/Channel ± 35V, POUT = 1W & 30W/Channel ± 25V, RL= 4Ω, 80kHz BW RL= 8Ω, 80kHz BW 200586e8 200586e5 9 www.national.com, THD+N vs Output Power/Channel THD+N vs Output Power/Channel ± 30V, RL= 6Ω, 80kHz BW ± 35V, RL= 8Ω, 80kHz BW 200586e9 200586f0 Output Power/Channel Output Power/Channel vs Supply Voltage vs Supply Voltage f = 1kHz, RL = 4Ω, 80kHz BW f = 1kHz, RL = 6Ω, 80kHzBW 20058625 20058626 www.national.com 10, Output Power/Channel Total Power Dissipation vs Supply Voltage vs Output Power/Channel f = 1kHz, RL = 8Ω, 80kHz BW 1% THD (max), RL = 4Ω, 80kHz BW 20058623 200586a6 Total Power Dissipation Total Power Dissipation vs Output Power/Channel vs Output Power/Channel 1% THD (max), RL = 6Ω, 80kHz BW 1% THD (max), RL = 8Ω, 80kHz BW 200586a7 200586a8 11 www.national.com, Crosstalk vs Frequency Crosstalk vs Frequency ± 25V, POUT = 10W ± 35V, POUT = 10W RL = 4Ω, 80kHz BW RL = 8Ω, 80kHz BW 200586c5 200586a5 Mute Attenuation Supply Current vs Mute Pin Current vs Supply Voltage POUT = 10W/Channel 200586b4 200586c6 Large Signal Response Power Supply Rejection Ratio 200586c7 200586c8 www.national.com 12, Common Mode Open Loop Rejection Ratio Frequency Response 200586d0 200586c9 Clipping Voltage Clipping Voltage vs Supply Voltage vs Supply Voltage 200586d1 200586d2 THD+N vs Frequency THD+N vs Frequency ± 25V, POUT = 1W & 50W ± 35V, POUT = 1W & 50W Bridge Mode (Note 18), RL = 8Ω, 80kHz BW Parallel Mode (Note 19), RL = 4Ω, 80kHz BW 200586e6 200586e7 13 www.national.com, THD+N vs Output Power THD+N vs Output Power ± 25V, Bridge Mode (Note 18) ± 35V, Parallel Mode (Note 19) RL = 8Ω, 80kHz BW RL = 4Ω, 80kHz BW 200586f1 200586f2 Output Power vs Output Power vs Supply Voltage, Bridge Mode (Note 18) Supply Voltage, Parallel Mode (Note 19) f = 1kHz, RL = 8Ω, 80kHz BW f = 1kHz, RL = 4Ω, 80kHz BW 20058627 20058624 Safe Area SPiKe Protection Response 200586d3 200586d4 www.national.com 14,

Frequency Response of Demo Board POUT = 10W/Channel = 0dB RIN = 47kΩ, RL = 8Ω, No Filters

200586e2 Note 18: Bridge mode graphs were taken using the demo board and inverting the signal to the channel B input. Note 19: Parallel mode graphs were taken using the demo board and connecting each output through a 0.1Ω/3W resistor to the load. 15 www.national.com,

Application Information supply voltages fully decay preventing transients on the out-

put. MUTE MODE OVER-VOLTAGE PROTECTION The muting function allows the user to mute the amplifier. This The LM4780 contains over-voltage protection circuitry that can be accomplished as shown in the Typical Application Cir- limits the output current while also providing voltage clamp- cuit. The resistor RM is chosen with reference to the negative ing. The clamp does not, however, use internal clamping supply voltage and is used in conjunction with a switch. The diodes. The clamping effect is quite the same because the switch, when opened or switched to GND, cuts off the current output transistors are designed to work alternately by sinking flow from the MUTE pins to −VEE, thus placing the LM4780 large current spikes. into mute mode. Refer to the Mute Attenuation vs Mute Cur- rent curves in the Typical Performance Characteristics SPiKe PROTECTION section for values of attenuation per current out of each MUTE The LM4780 is protected from instantaneous peak-tempera- pin. The resistance RM is calculated by the following equation: ture stressing of the power transistor array. The Safe Oper- ≤ ating graph in the Typical Performance CharacteristicsRM (|−VEE| − 2.6V) / IMUTE section shows the area of device operation where SPiKe Where I ≥ 0.5mA for each MUTE pin. Protection Circuitry is not enabled. The SPiKe Protection Re-MUTE sponse waveform graph shows the waveform distortion when The MUTE pins can be tied together so that only one resistor SPiKe is enabled. Please refer to AN-898 for more detailed is required for the mute function. The mute resistor value must information. be chosen so that a minimum of 1mA is pulled through the resistor RM. This ensures that each amplifier is fully opera- THERMAL PROTECTION tional. Taking into account supply line fluctuations, it is a good The LM4780 has a sophisticated thermal protection scheme idea to pull out 1mA per MUTE pin or 2mA total if both pins to prevent long-term thermal stress of the device. When the are tied together. temperature on the die exceeds 150°C, the LM4780 shuts A turn-on MUTE or soft start circuit may also be used during down. It starts operating again when the die temperature power up. A simple circuit like the one shown below may be drops to about 145°C, but if the temperature again begins to used. rise, shutdown will occur again above 150°C. Therefore, the device is allowed to heat up to a relatively high temperature if the fault condition is temporary, but a sustained fault will cause the device to cycle in a Schmitt Trigger fashion be- tween the thermal shutdown temperature limits of 150°C and 145°C. This greatly reduces the stress imposed on the IC by thermal cycling, which in turn improves its reliability under sustained fault conditions. Since the die temperature is directly dependent upon the heat sink used, the heat sink should be chosen so that thermal shutdown is not activated during normal operation. Using the 200586a3 best heat sink possible within the cost and space constraints The RC combination of CM and RM1 may cause the voltage at of the system will improve the long-term reliability of any pow- point A to change more slowly than the -VEE supply voltage. er semiconductor device, as discussed in the Determining Until the voltage at point A is low enough to have 0.5mA of the Correct Heat Sink section. current per MUTE pin flow through RM2, the IC will be in mute mode. The series combination of R and R needs to sat- DETERMlNlNG MAXIMUM POWER DISSIPATIONM1 M2 isfy the mute equation above for all operating voltages or mute Power dissipation within the integrated circuit package is a mode may be activated during normal operation. For a longer very important parameter requiring a thorough understanding turn-on mute time, a larger time constant, τ = RC = RM1CM if optimum power output is to be obtained. An incorrect max- (sec), is needed. For the values show above and with the imum power dissipation calculation may result in inadequate MUTE pins tied together, the LM4780 will enter play mode heat sinking causing thermal shutdown and thus limiting the when the voltage at point A is -17.6V. The voltage at point A output power. is found with Equation (1) below. Equation 2 shows the theoretical maximum power dissipation V (t) = (V - V )e-t/τ (Volts) (1) point of each amplifier in a single-ended configuration whereAfOVCC is the total supply voltage. where: t = time (sec)  τ PDMAX = (V22CC) / 2π RL (2) = RC (sec) Vo = Voltage on C at t = 0 (Volts) V = Final voltage, -V in this circuit (Volts) Thus by knowing the total supply voltage and rated outputf EE load, the maximum power dissipation point can be calculated. UNDER-VOLTAGE PROTECTION The package dissipation is twice the number which results Upon system power-up, the under-voltage protection circuitry from Equation 2 since there are two amplifiers in each allows the power supplies and their corresponding capacitors LM4780. Refer to the graphs of Power Dissipation versus to come up close to their full values before turning on the Output Power in the Typical Performance Characteristics LM4780. Since the supplies have essentially settled to their section which show the actual full range of power dissipation final value, no DC output spikes occur. At power down, the not just the maximum theoretical point that results from Equa- outputs of the LM4780 are forced to ground before the power tion 2. www.national.com 16, DETERMINING THE CORRECT HEAT SINK BRIDGED AMPLIFIER APPLICATION The choice of a heat sink for a high-power audio amplifier is The LM4780 has two operational amplifiers internally, allow- made entirely to keep the die temperature at a level such that ing for a few different amplifier configurations. One of these the thermal protection circuitry is not activated under normal configurations is referred to as “bridged mode” and involves circumstances. driving the load differentially through the LM4780's outputs. The thermal resistance from the die to the outside air, θ This configuration is shown in Figure 2. Bridged mode oper-JA (junction to ambient), is a combination of three thermal resis- ation is different from the classical single-ended amplifier tances, θ (junction to case), θ (case to sink), and θ (sink configuration where one side of its load is connected toJC CS SA to ambient). The thermal resistance, θ ground.JC (junction to case), of the LM4780T is 0.8°C/W. Using Thermalloy Thermacote ther- A bridge amplifier design has a distinct advantage over the mal compound, the thermal resistance, θCS (case to sink), is single-ended configuration, as it provides differential drive to about 0.2°C/W. Since convection heat flow (power dissipa- the load, thus doubling output swing for a specified supply tion) is analogous to current flow, thermal resistance is anal- voltage. Theoretically, four times the output power is possible ogous to electrical resistance, and temperature drops are as compared to a single-ended amplifier under the same con- analogous to voltage drops, the power dissipation out of the ditions. This increase in attainable output power assumes that LM4780 is equal to the following: the amplifier is not current limited or clipped. A direct consequence of the increased power delivered to the ) / θ load by a bridge amplifier is an increase in internal power dis-PDMAX = (TJMAX−TAMB JA (3) sipation. For each operational amplifier in a bridge configu- ration, the internal power dissipation will increase by a factor where TJMAX = 150°C, TAMB is the system ambient tempera- of two over the single ended dissipation. Thus, for an audio ture and θJA = θJC + θCS + θSA. power amplifier such as the LM4780, which has two opera- tional amplifiers in one package, the package dissipation will increase by a factor of four. To calculate the LM4780's max- imum power dissipation point for a bridged load, multiply Equation 2 by a factor of four. This value of PDMAX can be used to calculate the correct size heat sink for a bridged amplifier application. Since the internal dissipation for a given power supply and load is increased by using bridged-mode, the heatsink's θSA will have to decrease 20058652 accordingly as shown by Equation 4. Refer to the section, Once the maximum package power dissipation has been cal- Determining the Correct Heat Sink, for a more detailed culated using Equation 2, the maximum thermal resistance, discussion of proper heat sinking for a given application. θSA, (heat sink to ambient) in °C/W for a heat sink can be calculated. This calculation is made using Equation 4 which PARALLEL AMPLIFIER APPLICATION is derived by solving for θSA in Equation 3. Parallel configuration is normally used when higher output current is needed for driving lower impedance loads (i.e. 4Ω or lower) to obtain higher output power levels. As shown in θSA = [(TJMAX−TAMB)−PDMAX(θJC +θCS)] / PDMAX (4) Figure 3 , the parallel amplifier configuration consist of de- signing the amplifiers in the IC to have identical gain, con- Again it must be noted that the value of θ is dependent upon necting the inputs in parallel and then connecting the outputsSA the system designer's amplifier requirements. If the ambient in parallel through a small external output resistor. Any num- temperature that the audio amplifier is to be working under is ber of amplifiers can be connected in parallel to obtain the higher than 25°C, then the thermal resistance for the heat needed output current or to divide the power dissipation sink, given all other things are equal, will need to be smaller. across multiple IC packages. Ideally, each amplifier shares the output current equally. Due to slight differences in gain the SUPPLY BYPASSING current sharing will not be equal among all channels. If current The LM4780 has excellent power supply rejection and does is not shared equally among all channels then the power dis- not require a regulated supply. However, to improve system sipation will also not be equal among all channels. It is rec- performance as well as eliminate possible oscillations, the ommended that 0.1% tolerance resistors be used to set the LM4780 should have its supply leads bypassed with low-in- gain (Ri and Rf) for a minimal amount of difference in current ductance capacitors having short leads that are located close sharing. to the package terminals. Inadequate power supply bypass- When operating two or more amplifiers in parallel mode the ing will manifest itself by a low frequency oscillation known as impedance seen by each amplifier is equal to the total load “motorboating” or by high frequency instabilities. These in- impedance multiplied by the number of amplifiers driving the stabilities can be eliminated through multiple bypassing uti- load in parallel as shown by Equation 5 below: lizing a large tantalum or electrolytic capacitor (10μF or larger) which is used to absorb low frequency variations and a small ceramic capacitor (0.1μF) to prevent any high frequency feed- RL(parallel) = RL(total) * Number of amplifiers (5) back through the power supply lines. If adequate bypassing is not provided, the current in the sup- Once the impedance seen by each amplifier in the parallel ply leads which is a rectified component of the load current configuration is known then Equation (2) can be used with this may be fed back into internal circuitry. This signal causes dis- calculated impedance to find the amount of power dissipation tortion at high frequencies requiring that the supplies be by- for each amplifier. Total power dissipation (PDMAX) within an passed at the package terminals with an electrolytic capacitor IC package is found by adding up the power dissipation for of 470μF or more. each amplifier in the IC package. Using the calculated 17 www.national.com, PDMAX the correct heat sink size can be determined. Refer to V. Gain settings below 10V/V may experience instability and the section, Determining the Correct Heat Sink, for more using the LM4780 for gains higher than 50V/V will see an in- information and detailed discussion of proper heat sinking. crease in noise and THD. The combination of Ri with Ci (see Figure 1) creates a high SINGLE-SUPPLY AMPLIFIER APPLICATION pass filter. The low frequency response is determined by The typical application of the LM4780 is a split supply ampli- these two components. The -3dB point can be found from fier. But as shown in Figure 4, the LM4780 can also be used Equation 7 shown below: in a single power supply configuration. This involves using some external components to create a half-supply bias which is used as the reference for the inputs and outputs. Thus, the fi = 1 / (2πRiCi) (Hz) (7) signal will swing around half-supply much like it swings around ground in a split-supply application. Along with proper If an input coupling capacitor is used to block DC from the circuit biasing, a few other considerations must be accounted inputs as shown in Figure 5, there will be another high pass for to take advantage of all of the LM4780 functions, like the filter created with the combination of CIN and RIN. When using mute function. a input coupling capacitor RIN is needed to set the DC bias point on the amplifier's input terminal. The resulting -3dB fre- CLICKS AND POPS quency response due to the combination of C and R can In the typical application of the LM4780 as a split-supply audio IN INbe found from Equation 8 shown below: power amplifier, the IC exhibits excellent “click” and “pop” performance when utilizing the mute mode. In addition, the device employs Under-Voltage Protection, which eliminates fIN = 1 / (2πRINCIN) (Hz) (8) unwanted power-up and power-down transients. The basis for these functions are a stable and constant half-supply po- tential. In a split-supply application, ground is the stable half- With large values of RIN oscillations may be observed on the supply potential. But in a single-supply application, the half- outputs when the inputs are left floating. Decreasing the value supply needs to charge up at the same rate as the supply rail, of RIN or not letting the inputs float will remove the oscillations. VCC. This makes the task of attaining a clickless and popless If the value of RIN is decreased then the value of CIN will need turn-on more challenging. Any uneven charging of the ampli- to increase in order to maintain the same -3dB frequency re- fier inputs will result in output clicks and pops due to the sponse. differential input topology of the LM4780. HIGH PERFORMANCE CONSIDERATIONS To achieve a transient free power-up and power-down, the Using low cost electrolytic capacitors in the signal path such voltage seen at the input terminals should be ideally the same. as C and C (see Figures 1 - 5) will result in very good per- Such a signal will be common-mode in nature, and will be IN iformance. However, electrolytic capacitors are less linear rejected by the LM4780. In Figure 4, the resistor RINP serves than other premium capacitors. Higher THD+N performance to keep the inputs at the same potential by limiting the voltage may be obtained by using high quality polypropylene capac- difference possible between the two nodes. This should sig- itors in the signal path. A more cost effective solution may be nificantly reduce any type of turn-on pop, due to an uneven the use of smaller value premium capacitors in parallel with charging of the amplifier inputs. This charging is based on a the larger electrolytic capacitors. This will maintain signal specific application loading and thus, the system designer quality in the upper audio band where any degradation is most may need to adjust these values for optimal performance. noticeable while also coupling in the signals in the lower audio As shown in Figure 4, the resistors labeled RBI help bias up band for good bass response. the LM4780 off the half-supply node at the emitter of the Distortion is introduced as the audio signal approaches the 2N3904. But due to the input and output coupling capacitors lower -3dB point, determined as discussed in the section in the circuit, along with the negative feedback, there are twoΩΩabove. By using larger values of capacitors such that the -3dBdifferent values of RBI, namely 10k and 200k . These re- point is well outside of the audio band will reduce this distor- sistors bring up the inputs at the same rate resulting in a tion and improve THD+N performance. popless turn-on. Adjusting these resistors values slightly may reduce pops resulting from power supplies that ramp ex- Increasing the value of the large supply bypass capacitors will tremely quick or exhibit overshoot during system turn-on. improve burst power output. The larger the supply bypass capacitors the higher the output pulse current without supply PROPER SELECTION OF EXTERNAL COMPONENTS droop increasing the peak output power. This will also in- Proper selection of external components is required to meet crease the headroom of the amplifier and reduce THD. the design targets of an application. The choice of external SIGNAL-TO-NOISE RATIO component values that will affect gain and low frequency re- sponse are discussed below. In the measurement of the signal-to-noise ratio, misinterpre- tations of the numbers actually measured are common. One The gain of each amplifier is set by resistors Rf and Ri for the amplifier may sound much quieter than another, but due to non-inverting configuration shown in Figure 1. The gain is improper testing techniques, they appear equal in measure- found by Equation 6 below: ments. This is often the case when comparing integrated circuit designs to discrete amplifier designs. Discrete transis- AV = 1 + Rf / Ri (V/V) (6) tor amps often “run out of gain” at high frequencies and therefore have small bandwidths to noise as indicated below. For best noise performance, lower values of resistors are used. A value of 1kΩ is commonly used for Ri and then setting the value of Rf for the desired gain. For the LM4780 the gain should be set no lower than 10V/V and no higher than 50V/ www.national.com 18, with the heat sink will increase the thermal resistance from the package case to the heat sink (θCS) resulting in higher operating die temperatures and possible unwanted thermal shut down activation. Extreme over tightening of the mounting screws will cause severe physical stress resulting in cracked die and catastrophic IC failure. The recommended mounting screw size is M3 with a maximum torque of 50 N-cm. Addi- tionally, it is best to use washers under the screws to distribute the force over a wider area or a screw with a wide flat head. To further distribute the mounting force a solid mounting bar in front of the package and secured in place with the two mounting screws may be used. Other mounting options in- Integrated circuits have additional open loop gain allowing clude a spring clip. If the package is secured with pressure on additional feedback loop gain in order to lower harmonic dis- the front of the package the maximum pressure on the molded tortion and improve frequency response. It is this additional plastic should not exceed 150N/mm 2. bandwidth that can lead to erroneous signal-to-noise mea- Additionally, if the mounting screws are used to force the surements if not considered during the measurement pro- package into correct alignment with the heat sink, package cess. In the typical example above, the difference in stress will be increased. This increase in package stress will bandwidth appears small on a log scale but the factor of 10in result in reduced contact area with the heat sink increasing bandwidth, (200kHz to 2MHz) can result in a 10dB theoretical die operating temperature and possible catastrophic IC fail- difference in the signal-to-noise ratio (white noise is propor- ure. tional to the square root of the bandwidth in a system). LAYOUT, GROUND LOOPS AND STABILITY In comparing audio amplifiers it is necessary to measure the magnitude of noise in the audible bandwidth by using a The LM4780 is designed to be stable when operated at a “weighting” filter (Note 16). A “weighting” filter alters the fre- closed-loop gain of 10 or greater, but as with any other high- quency response in order to compensate for the average current amplifier, the LM4780 can be made to oscillate under human ear's sensitivity to the frequency spectra. The weight- certain conditions. These oscillations usually involve printed ing filters at the same time provide the bandwidth limiting as circuit board layout or output/input coupling issues. discussed in the previous paragraph. When designing a layout, it is important to return the load In addition to noise filtering, differing meter types give different ground, the output compensation ground, and the low level noise readings. Meter responses include: (feedback and input) grounds to the circuit board common ground point through separate paths. Otherwise, large cur- 1. RMS reading, rents flowing along a ground conductor will generate voltages 2. average responding, on the conductor which can effectively act as signals at the 3. peak reading, and input, resulting in high frequency oscillation or excessive dis- 4. quasi peak reading. tortion. It is advisable to keep the output compensation com- Although theoretical noise analysis is derived using true RMS ponents and the 0.1μF supply decoupling capacitors as close based calculations, most actual measurements are taken with as possible to the LM4780 to reduce the effects of PCB trace ARM (Average Responding Meter) test equipment. resistance and inductance. For the same reason, the ground return paths should be as short as possible. Typical signal-to-noise figures are listed for an A-weighted fil- ter which is commonly used in the measurement of noise. The In general, with fast, high-current circuitry, all sorts of prob- shape of all weighting filters is similar, with the peak of the lems can arise from improper grounding which again can be curve usually occurring in the 3kHz–7kHz region. avoided by returning all grounds separately to a common point. Without isolating the ground signals and returning the LEAD INDUCTANCE grounds to a common point, ground loops may occur. Power op amps are sensitive to inductance in the output “Ground Loop” is the term used to describe situations occur- leads, particularly with heavy capacitive loading. Feedback to ring in ground systems where a difference in potential exists the input should be taken directly from the output terminal, between two ground points. Ideally a ground is a ground, but minimizing common inductance with the load. unfortunately, in order for this to be true, ground conductors Lead inductance can also cause voltage surges on the sup- with zero resistance are necessary. Since real world ground plies. With long leads to the power supply, energy is stored in leads possess finite resistance, currents running through the lead inductance when the output is shorted. This energy them will cause finite voltage drops to exist. If two ground re- can be dumped back into the supply bypass capacitors when turn lines tie into the same path at different points there will the short is removed. The magnitude of this transient is re- be a voltage drop between them. The first figure below shows duced by increasing the size of the bypass capacitor near the a common ground example where the positive input ground IC. With at least a 20μF local bypass, these voltage surges and the load ground are returned to the supply ground point are important only if the lead length exceeds a couple feet via the same wire. The addition of the finite wire resistance, (>1μH lead inductance). Twisting together the supply and R2, results in a voltage difference between the two points as ground leads minimizes the effect. shown below. PHYSICAL IC MOUNTING CONSIDERATIONS Mounting of the package to a heat sink must be done such that there is sufficient pressure from the mounting screws to insure good contact with the heat sink for efficient heat flow. Over tightening the mounting screws will cause the package to warp reducing contact area with the heat sink. Less contact 19 www.national.com, single-point grounding are most common among printed cir- cuit board designs, since the circuit is surrounded by large ground areas which invite the temptation to run a device to the closest ground spot. As a final rule, make all ground re- turns low resistance and low inductance by using large wire and wide traces. Occasionally, current in the output leads (which function as antennas) can be coupled through the air to the amplifier in- put, resulting in high-frequency oscillation. This normally hap- pens when the source impedance is high or the input leads are long. The problem can be eliminated by placing a small capacitor, CC, (on the order of 50pF to 500pF) across the LM4780 input terminals. Refer to the External Components Description section relating to component interaction with Cf. REACTIVE LOADING It is hard for most power amplifiers to drive highly capacitive loads very effectively and normally results in oscillations or ringing on the square wave response. If the output of the LM4780 is connected directly to a capacitor with no series resistance, the square wave response will exhibit ringing if the capacitance is greater than about 0.2μF. If highly capacitive loads are expected due to long speaker cables, a method commonly employed to protect amplifiers from low impedances at high frequencies is to couple to the load through a 10Ω resistor in parallel with a 0.7μH inductor. The inductor-resistor combination as shown in the Figure 5 iso- lates the feedback amplifier from the load by providing high 20058698 output impedance at high frequencies thus allowing the 10Ω The load current IL will be much larger than input bias current resistor to decouple the capacitive load and reduce the Q of II, thus V1 will follow the output voltage directly, i.e. in phase. the series resonant circuit. The LR combination also provides Therefore the voltage appearing at the non-inverting input is low output impedance at low frequencies thus shorting out the effectively positive feedback and the circuit may oscillate. If 10Ω resistor and allowing the amplifier to drive the series RC there was only one device to worry about then the values of load (large capacitive load due to long speaker cables) di- R1 and R2 would probably be small enough to be ignored; rectly. however, several devices normally comprise a total system. Any ground return of a separate device, whose output is in INVERTING AMPLIFIER APPLICATION phase, can feedback in a similar manner and cause instabil- The inverting amplifier configuration may be used instead of ities. Out of phase ground loops also are troublesome, caus- the more common non-inverting amplifier configuration ing unexpected gain and phase errors. shown in Figure 1. The inverting amplifier can have better The solution to most ground loop problems is to always use THD+N performance and eliminates the need for a large ca- a single-point ground system, although this is sometimes im- pacitor (Ci) reducing cost and space requirements. The val- practical. The third figure above is an example of a single- ues show in Figure 6 are only one example of an amplifier point ground system. with a gain of 20V/V (Gain = -Rf/Ri). For different resistor val- The single-point ground concept should be applied rigorously ues, the value of RB should be eqaul to the parallel combina- to all components and all circuits when possible. Violations of tion of Rf and Ri. www.national.com 20, FIGURE 6. Inverting Amplifier Application Circuit 21 www.national.com, 200586f3 FIGURE 7. Reference PCB Schematic www.national.com 22,

LM4780 REFERENCE BOARD ARTWORK

200586d9 200586d8

Composite Layer Silk Layer

200586d7 200586e0

Top Layer Bottom Layer

23 www.national.com,

Bill Of Materials For Reference Pcb

Symbol Value Tolerance Type/Description Comment RIN1, RIN2 15kΩ 5% 1/4 Watt RB1, RB2 1kΩ 1% 1/4 Watt RF1, RF2 20kΩ 1% 1/4 Watt Ri1, Ri2 1kΩ 1% 1/4 Watt RSN1, RSN2, 2.7Ω 5% 1/4 Watt RG 2.7Ω 5% 1/4 Watt RM 10kΩ 5% 1/4 Watt CIN1, CIN2 1µF 10% Metallized Polyester Film Ci1, Ci2, 68µF 20% Electrolytic Radial / 50V CSN1, CSN2 0.1µF 20% Monolithic Ceramic CN1, CN2 15pF 20% Monolithic Ceramic CS1, CS2, CS3 0.1µF 20% Monolithic Ceramic CS4, CS5, CS6 10µF 20% Electrolytic Radial / 50V CS7, CS8 1,000µF 20% Electrolytic Radial / 50V S1 SPDT (on-on) Switch Non-Switched PC Mount RCA J1, J2 Jack J4, J7, J8 PCB Banana Jack - BLACK J3, J5, J6, J9 PCB Banana Jack - RED 27 lead TO-220 Power Socket U1 with push lever release or LM4780

IC

www.national.com 24,

Physical Dimensions inches (millimeters) unless otherwise noted

Non-Isolated TO-220 27-Lead Package Order Number LM4780TA NS Package Number TA27A 25 www.national.com,

Notes

For more National Semiconductor product information and proven design tools, visit the following Web sites at: www.national.com Products Design Support Amplifiers www.national.com/amplifiers WEBENCH® Tools www.national.com/webench Audio www.national.com/audio App Notes www.national.com/appnotes Clock and Timing www.national.com/timing Reference Designs www.national.com/refdesigns Data Converters www.national.com/adc Samples www.national.com/samples Interface www.national.com/interface Eval Boards www.national.com/evalboards LVDS www.national.com/lvds Packaging www.national.com/packaging Power Management www.national.com/power Green Compliance www.national.com/quality/green Switching Regulators www.national.com/switchers Distributors www.national.com/contacts LDOs www.national.com/ldo Quality and Reliability www.national.com/quality LED Lighting www.national.com/led Feedback/Support www.national.com/feedback Voltage References www.national.com/vref Design Made Easy www.national.com/easy PowerWise® Solutions www.national.com/powerwise Applications & Markets www.national.com/solutions Serial Digital Interface (SDI) www.national.com/sdi Mil/Aero www.national.com/milaero Temperature Sensors www.national.com/tempsensors SolarMagic™ www.national.com/solarmagic PLL/VCO www.national.com/wireless PowerWise® Design www.national.com/training University THE CONTENTS OF THIS DOCUMENT ARE PROVIDED IN CONNECTION WITH NATIONAL SEMICONDUCTOR CORPORATION (“NATIONAL”) PRODUCTS. NATIONAL MAKES NO REPRESENTATIONS OR WARRANTIES WITH RESPECT TO THE ACCURACY OR COMPLETENESS OF THE CONTENTS OF THIS PUBLICATION AND RESERVES THE RIGHT TO MAKE CHANGES TO SPECIFICATIONS AND PRODUCT DESCRIPTIONS AT ANY TIME WITHOUT NOTICE. NO LICENSE, WHETHER EXPRESS, IMPLIED, ARISING BY ESTOPPEL OR OTHERWISE, TO ANY INTELLECTUAL PROPERTY RIGHTS IS GRANTED BY THIS DOCUMENT. TESTING AND OTHER QUALITY CONTROLS ARE USED TO THE EXTENT NATIONAL DEEMS NECESSARY TO SUPPORT NATIONAL’S PRODUCT WARRANTY. EXCEPT WHERE MANDATED BY GOVERNMENT REQUIREMENTS, TESTING OF ALL PARAMETERS OF EACH PRODUCT IS NOT NECESSARILY PERFORMED. NATIONAL ASSUMES NO LIABILITY FOR APPLICATIONS ASSISTANCE OR BUYER PRODUCT DESIGN. BUYERS ARE RESPONSIBLE FOR THEIR PRODUCTS AND APPLICATIONS USING NATIONAL COMPONENTS. PRIOR TO USING OR DISTRIBUTING ANY PRODUCTS THAT INCLUDE NATIONAL COMPONENTS, BUYERS SHOULD PROVIDE ADEQUATE DESIGN, TESTING AND OPERATING SAFEGUARDS. EXCEPT AS PROVIDED IN NATIONAL’S TERMS AND CONDITIONS OF SALE FOR SUCH PRODUCTS, NATIONAL ASSUMES NO LIABILITY WHATSOEVER, AND NATIONAL DISCLAIMS ANY EXPRESS OR IMPLIED WARRANTY RELATING TO THE SALE AND/OR USE OF NATIONAL PRODUCTS INCLUDING LIABILITY OR WARRANTIES RELATING TO FITNESS FOR A PARTICULAR PURPOSE, MERCHANTABILITY, OR INFRINGEMENT OF ANY PATENT, COPYRIGHT OR OTHER INTELLECTUAL PROPERTY RIGHT. LIFE SUPPORT POLICY NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS PRIOR WRITTEN APPROVAL OF THE CHIEF EXECUTIVE OFFICER AND GENERAL COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein: Life support devices or systems are devices which (a) are intended for surgical implant into the body, or (b) support or sustain life and whose failure to perform when properly used in accordance with instructions for use provided in the labeling can be reasonably expected to result in a significant injury to the user. A critical component is any component in a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system or to affect its safety or effectiveness. National Semiconductor and the National Semiconductor logo are registered trademarks of National Semiconductor Corporation. All other brand or product names may be trademarks or registered trademarks of their respective holders. Copyright© 2010 National Semiconductor Corporation For the most current product information visit us at www.national.com National Semiconductor National Semiconductor Europe National Semiconductor Asia National Semiconductor Japan Americas Technical Technical Support Center Pacific Technical Support Center Technical Support Center Support Center Email: email is hidden Email: email is hidden Email: email is hidden Email: email is hidden Tel: 1-800-272-9959 www.national.com

LM4780 Overture® Audio Power Amplifier Series Stereo 60W, Mono 120W Audio Power Amplifier with Mute

,

IMPORTANT NOTICE

Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, modifications, enhancements, improvements, and other changes to its products and services at any time and to discontinue any product or service without notice. Customers should obtain the latest relevant information before placing orders and should verify that such information is current and complete. All products are sold subject to TI’s terms and conditions of sale supplied at the time of order acknowledgment. TI warrants performance of its hardware products to the specifications applicable at the time of sale in accordance with TI’s standard warranty. Testing and other quality control techniques are used to the extent TI deems necessary to support this warranty. Except where mandated by government requirements, testing of all parameters of each product is not necessarily performed. TI assumes no liability for applications assistance or customer product design. Customers are responsible for their products and applications using TI components. To minimize the risks associated with customer products and applications, customers should provide adequate design and operating safeguards. TI does not warrant or represent that any license, either express or implied, is granted under any TI patent right, copyright, mask work right, or other TI intellectual property right relating to any combination, machine, or process in which TI products or services are used. Information published by TI regarding third-party products or services does not constitute a license from TI to use such products or services or a warranty or endorsement thereof. Use of such information may require a license from a third party under the patents or other intellectual property of the third party, or a license from TI under the patents or other intellectual property of TI. Reproduction of TI information in TI data books or data sheets is permissible only if reproduction is without alteration and is accompanied by all associated warranties, conditions, limitations, and notices. Reproduction of this information with alteration is an unfair and deceptive business practice. TI is not responsible or liable for such altered documentation. Information of third parties may be subject to additional restrictions. Resale of TI products or services with statements different from or beyond the parameters stated by TI for that product or service voids all express and any implied warranties for the associated TI product or service and is an unfair and deceptive business practice. TI is not responsible or liable for any such statements. TI products are not authorized for use in safety-critical applications (such as life support) where a failure of the TI product would reasonably be expected to cause severe personal injury or death, unless officers of the parties have executed an agreement specifically governing such use. Buyers represent that they have all necessary expertise in the safety and regulatory ramifications of their applications, and acknowledge and agree that they are solely responsible for all legal, regulatory and safety-related requirements concerning their products and any use of TI products in such safety-critical applications, notwithstanding any applications-related information or support that may be provided by TI. Further, Buyers must fully indemnify TI and its representatives against any damages arising out of the use of TI products in such safety-critical applications. TI products are neither designed nor intended for use in military/aerospace applications or environments unless the TI products are specifically designated by TI as military-grade or "enhanced plastic." Only products designated by TI as military-grade meet military specifications. Buyers acknowledge and agree that any such use of TI products which TI has not designated as military-grade is solely at the Buyer's risk, and that they are solely responsible for compliance with all legal and regulatory requirements in connection with such use. TI products are neither designed nor intended for use in automotive applications or environments unless the specific TI products are designated by TI as compliant with ISO/TS 16949 requirements. Buyers acknowledge and agree that, if they use any non-designated products in automotive applications, TI will not be responsible for any failure to meet such requirements. Following are URLs where you can obtain information on other Texas Instruments products and application solutions: Products Applications Audio www.ti.com/audio Communications and Telecom www.ti.com/communications Amplifiers amplifier.ti.com Computers and Peripherals www.ti.com/computers Data Converters dataconverter.ti.com Consumer Electronics www.ti.com/consumer-apps DLP® Products www.dlp.com Energy and Lighting www.ti.com/energy DSP dsp.ti.com Industrial www.ti.com/industrial Clocks and Timers www.ti.com/clocks Medical www.ti.com/medical Interface interface.ti.com Security www.ti.com/security Logic logic.ti.com Space, Avionics and Defense www.ti.com/space-avionics-defense Power Mgmt power.ti.com Transportation and Automotive www.ti.com/automotive Microcontrollers microcontroller.ti.com Video and Imaging www.ti.com/video RFID www.ti-rfid.com OMAP Mobile Processors www.ti.com/omap Wireless Connectivity www.ti.com/wirelessconnectivity TI E2E Community Home Page e2e.ti.com Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265 Copyright © 2011, Texas Instruments Incorporated]
15

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