Download: APPLICATION NOTE 100 − 450 MHz 250 W Power Amplifier with the BLF548 MOSFET AN98021

APPLICATION NOTE 100 − 450 MHz 250 W Power Amplifier with the BLF548

MOSFET

AN98021,

CONTENTS

1 INTRODUCTION 2 DESIGN CONSIDERATIONS 3 AMPLIFIER CONCEPT 4 INPUT CIRCUITRY 5 ADJUSTMENT OF THE AMPLIFIER 5.1 Tuning the outputnetwork 5.2 Testing the unit under RF conditions 5.3 Tuning the unit’s inputnetwork 5.4 Combining the units 6 CONCLUSIONS 7 REFERENCES 8 APPENDIXA9APPENDIX B 10 APPENDIX C 1998 Mar 23 2, 1 INTRODUCTION In this report the design procedures and measurement results are given of a two octave wideband amplifier (covering both the civil and military airbands between 100 and 450 MHz), equiped with two MOSFET devices, which is capable of generating 250 W of output power. In order to achieve a good broadband capability one has to use devices with the output capacitance reduced to the utmost minimum. While applying PHILIPS’ BLF548 MOSFETs it was possible to obtain a respectable powergain of more than 10 dB, throughout the whole band. The BLF548 is a balanced N-channel enhancement mode vertical D-MOS transistor in a SOT262 package, especially designed for use in wideband amplifiers up to 500 MHz. The transistor is capable to deliver 150 W nominal outputpower at a supply voltage of 28 Volts. Due to the low output capacitance the attainable bandwidth will exceed 300 MHz. 2 DESIGN CONSIDERATIONS While designing broadband amplifiers, one has to take several things into account: • To select the right manufacturer, able to supply the products with a good reliability, gives a good support and offers a complete range of transistors e.g. for driverstages. • To select the right active components, capable to fulfill the desired wishes, such as; high reliability, high powergain, high efficiency, excellent mismatch capabilities, right loadpower, good long-life properties and last but not least; good broadband capability. • To terminate the transistor with the right load impedance, with other words, to determinate the right output matching network. • To eliminate the 6 dB/octave gain slope throughout the band of operation, in order to achieve an acceptable gainflatness. • To find the right input matching network; the input VSWR has to be low in order to achieve a good termination for the driverstage. • To design the matching networks in such a way that they are capable to handle the, at some points very high, R.F. currents. A balanced transistor was chosen in order to reduce the second harmonic (due to the push-pull effect) and to reduce the number of required components. The criteria for chosen MOSFETs over bipolar transistors are; high powergain, high load mismatch capabilities, low noise and easy biasing. Nowadays three major MOSFET suppliers are involved when Pl = 150 W is needed at f = 500 MHz and Vds = 28 V. Available are; BLF548, industry type A and industry type B. Table 1 gives an overview of the characteristics of these 3 types. Table 1 BLF548 TYPE A TYPE B UNIT f 500 400 500 MHz Gp >10 >10 >8 dB ηd >50 >50 >55 % Ciss 105 180 140 pF Coss 90 200 100 pF Crss 25 20 32 pF BW 300 133 266 MHz 1998 Mar 23 3, With: BW = 1/(2π * Rload * Co); Rload = Vds2/(2 * Pl) and Co = 1.15 * Coss. BW = bandwidth, Rload = loadresistance, Co = outputcapacitance. In order to achieve the best possible broadband results, the BLF548 is a very good choice. Other Philips MOSFETs in the 500 MHz series are, followed by nominal loadpower: 12.5 Volts − single ended BLF5212WBLF5225W28 Volts − single ended BLF5425WBLF543 10 W BLF544 20 W 28 Volts − push-pull BLF544B 20 W BLF545 40 W BLF546 80 W BLF547 100 W BLF548 150W3AMPLIFIER CONCEPT The amplifier concept described in this paper is based upon two identical modular units, each containing one BLF548 MOSFET. Both units are combined by means of two 3 dB − 90° hybrid couplers, which is shown in Fig.1. The main advantage is that the input VSWR will be very good; since it is independent of the mismatch introduced by the units, the 50 Ω termination will cause a good load for the driver stage, e.g. equipped with BLF544. 1998 Mar 23 4, handbook, full pagewidth UNIT 1 1:4 1:4 BALUN BLF548 BALUN 1:4 1:4 50 Ω input 50 Ω output 3 dB 3 dB 90° 90° UNIT 2 HYBRID HYBRID 1:4 1:4 BALUN BLF548 BALUN MGH784 1:4 1:4 Fig.1 Schematical representation of the concept of the eventual amplifier. The following seven steps have been followed in order to develop a first prototype of one unit. 1. Determine the BLF548’s 150 W output power load impedance between 100 and 500 MHz (50 MHz interval steps) by measurement techniques or simulations. At the moment of writing it was not possible to perform full automatic measurements at frequencies lower than 500 MHz with transistors build in a balanced SOT262 header. Therefore the load impedances have been calculated by means of the electrical equivalent diagram shown in Fig.2. 2. Find the correct output matching network which transforms the 50 Ω termination to the required load impedance for the frequency range 100-500 MHz. 3. Optimize the outputmatching network of step 2 with help of linear simulation software, such as Touchstone (EESOF). 4. Since the matching network will not have an ideal behaviour, it is necessary to determine the actual load impedance of the selected output matching network, again in 50 MHz steps between 100-500 MHz. 5. Calculate (or even better, determine by means of load-pull measurements) both the powergain and input impedance of the transistor by presenting the load impedences, found at step 4, to it. This is very important to investigate the behaviour of the transistor while terminating it with the selected output matching network. 6. Choose the right input matching network which has a minimum returnloss (Rl) at the highest frequency (450 MHz) and a declining Rl for lower frequencies in a way that the gain increase effect for lower frequencies is equalized. Other possibilities, as feedback or frequency dependent damping at the gate side (by means of low Rgs), can be taken into consideration. 7. Optimize the input network for gainflatness by means of linear simulation software (Touchstone, EESOF). Remember the input VSWR throughout the band is taken care of by the use of 90° hybrids, which combine the two modular units. 1998 Mar 23 5, handbook, full pagewidth L Cg Rg gd Rd Ld gate drain Cgs Cds GFS R Cs s Ls source source MGH785 Fig.2 RF Power MOSFET equivalent diagram (one BLF548 section). At the following pages the design steps are presented which were followed at PHILIPS’ laboratories in order to design a 150 W unit. Using the diagram shown in Fig.2, powergain and impedances have been calculated first, using the data given in Table 2. Table 2 Lg 0.58 nH Ls 0.11 nH Ld 0.50 nH Rg 0.09 Ω Rs 0.08 Ω Rd 0.19 Ω Cgd 29 pF 1.15 × Crss Cgs 120 pF 1.5 × (Ciss-Crss) Cds 72 pF 1.15 × (Coss-Crss-Cs) Cs 2.4 pF 2.4 pF Gfs’ 1.6 S 0.5 × Gfs (for Class B) Rg, Rd, Rs are derived from Rdson measurements, Gfs and Cs are measured, Cgs, Cds, Cgd derived from measured Ciss, Coss, Crss respectively. Lg, Ls and Ld are calculated. Some of the assumptions are based on empirical rules and have proven to be correct in the past. Gp and Zin can now be calculated: 1998 Mar 23 6, Zin = Ri + jXi Gp = 10 * 10log (Gfs’ × Rload/ω2 × Ls × Ci) with; Xi = ω × Li − 1/(ω × Ci) Ri = (Gfs’ × Ls)/Ci Li = Lq + (Ls × Cgs)/Ci Ci = Cgs + Cgd (1 + Gfs’ × Rload) w = 2 πf Zload is chosen for maximum broadband capability. Table 3 Calculated powergain, Zin and required Zload (series components) F (MHz) PL (W) Gp (dB) ZIN (Ω) ZLOAD (Ω) 100 78.8 26.7 0.43 − j4.1 4.7 + j1.5 150 78.8 23.3 0.43 − j2.5 4.0 + j1.0 200 78.8 20.8 0.42 − j1.7 3.4 + j2.0 250 78.8 18.4 0.43 − j1.1 2.8 + j1.9 300 78.8 17.2 0.43 − j0.7 2.3 + j1.7 350 78.8 15.8 0.43 − j.0.3 1.9 + j1.4 400 78.8 14.5 0.44 − j0.0 1.6 + j1.1 450 78.8 13.4 0.44 + j0.2 1.3 + j0.7 500 78.8 12.4 0.45 + j0.5 1.1 + j0.4 The data is also given in datahandbook “RF power MOS transistors” − Philips Components. It can be noticed that without any gaincompensation the powergain difference between 100 and 500 MHz will exceed 10 dB. To terminate the transistor with the required loadimpedance, with respect to the broadband capability, the unbalanced 50 Ω load has to be transformed as close as possible to the loadimpedance as shown in Table 3. (Note: the impedances shown are based on one section, since the transistor is of a balanced type, Zin and Zload are related to virtual ground). To reduce the number of components which would be needed in case of a lumped element solution, a coaxial semi-rigid balun is used to transform the unbalanced 50 Ω load into two 25 Ω sections that are 180° apart in phase and 90° away from virtual ground. This is followed by a coaxial 4 : 1 transformer, with a characteristic impedance of 25 Ω. The result of this is: Rp = (√25 × 25)/4 = 6.2 Ω, which is close to the required Rload of the transistor. In order to give a good description of the outputnetwork, it will be described as a 3-port: one port terminated with 50 Ω unbalanced, the other two terminated with the transistor’s outputimpedance (the complex conjugate of loadimpedance). A computer listing of the outputnetwork is given in “Appendix A”. After optimizing the network to minimum returnloss (S11), while checking S13, the optimized return loss (in dB) of this network has been determined, see Fig.3. As a next step now the difference between the required and the network related loadimpedance can be (re-) calculated. The result on powergain (Gp) and imputimpedance (Zin) is given in Table 4. 1998 Mar 23 7, Table 4 Result on Zin and Gp as a result of the presented outputmatching network F (MHz) PL (W) Gp (dB) ZIN (Ω) ZLOAD (Ω) 100 78.6 32.7 0.11 − j4.1 4.3 + j1.0 150 78.8 19.7 1.04 − j2.9 4.1 + j0.2 200 78.8 17.3 1.02 − j2.2 3.3 + j0.1 250 78.8 16.0 0.87 − j1.5 2.9 + j0.1 300 78.8 14.9 0.79 − j0.9 2.8 + j0.3 350 78.8 13.5 0.80 − j.0.4 2.8 + j0.4 400 78.8 12.4 0.79 − j0.1 2.5 + j0.2 450 78.8 12.0 0.67 + j0.1 1.9 + j0.1 500 72.0 12.4 0.43 + j0.5 1.1 + j0.6 MGH786 handbook, halfpage −10 −20 −30 50 150 250 350 450 550 frequency (MHz) Fig.3 Simulated network response (output side). 4 INPUT CIRCUITRY Since Zin and Gp are now determined in a accurate way, the inputcircuitry can be determined. Special attention is given to the flatness of the gain as a function of frequency. The input network also consists of a coaxial balum, followed bya1: 4 coaxial transformer, both made of semi-rigid coaxial cable. Since Zin is rather low the characteristic impedance of the 1 : 4 transformer was chosen to be 10 Ω. The result of this is: Rp = (√25 × 10)/4 = 3.9 Ω. 1998 Mar 23 8, In order to compensate for the 6 dB/octave slope, matching to Zin is achieved at 450 MHz. At lower frequencies a mismatch is created, resulting in a decrease of powergain inversely proportional to the increase of the gain related to the transistor’s 6 dB/octave slope. The network listing of the input circuitry, again presented as a 3-port, is given in “Appendix B”. The network response (both input returnloss and predicted powergain) is given in Fig.4. Finally the schematic diagram and list of components are given in Fig.5. The unit’s layout is given in Fig.6. Note: two toroidal cores around T2 and T3 are used to prevent oscillations. 5 ADJUSTMENT OF THE AMPLIFIER 5.1 Tuning the outputnetwork In order to terminate the transistor with the proper load impedance, first the output network has to be tuned. The transistor was replaced by a dummyload, representing the transistors output impedance under full power conditions. The dummyload was realized after fitting the data of Table 3. To the dummyload model (roughly Rload in parallel with Coss, in series with draininductance Ld). Later the model was compensated for parasitics of both SOT262 header and network components. Initial settings for each side of the dummyload are: Rload = Vds2/2 × Pl = 5.2ΩC= 1.15 × Coss = 104 pF L = Ld = 0.5 nH The network listing is given in “Appendix C”. The final result, the dummyload lay-out, is given in Fig.7. MGH787 handbook, halfpage (1) (2) −10 −20 100 200 300 400 500 frequency (MHz) (1) DB [S12]. (2) DB [S22]. Fig.4 Simulated network response of inputside (predicted Gp = f(f)). 1998 Mar 23 9, handbook, full pagewidth L10 V L11BIAS C12 C17 R1 R2

V

R8 drain C1 C5 C6 R5 L9 C18 R3 R4 C2 T2 T4 C15BLF548 L1 L2 L5L6 T1 T6 50 Ω input C7 50 ΩC10 C11 C14 output C3 T3 L3 L4 T5 C16 C19L7 L8 MGH788 C4 R5 C13 R7 C8 C9 Fig.5 Schematical diagram and list of components of one unit. 1998 Mar 23 10, List of components DESIGNATION DESCRIPTION VALUE DIMENSIONS CATALOGUE NO. C1, C17 multilayer ceramic chip capacitor 100 nF 2222 852 47104 C2, C3 multilayer ceramic chip capacitor (note 1) 47 pF C4, C5, C8 multilayer ceramic chip capacitor (note 1) 820 pF C6, C9 multilayer ceramic chip capacitor (note 1) 300 pF C7 film dielectric trimmer 2-18 pF 2222 809 09006 C10, C14 film dielectric trimmer 2-9 pF 2222 809 09005 C11 multilayer ceramic chip capacitor (note 2) 39 pF C12 capacitor 22 nF C13 capacitor 100 nF C15, C16 multilayer ceramic chip capacitor (note 1) 120 pF C18 63 V electorlytic capacitor 1 µF 2222 685 78108 C19 film dielectric trimmer 1-5 pF 222 808 09004 L1, L3 stripline (note 3) 20Ω5× 8 mm L2, L4 stripline (note 3) 20 Ω 2.5 × 8 mm L5, L7 stripline (note 3) 20 Ω 11.5 × 8 mm L6, L8 stripline (note 3) 20Ω4× 8 mm L9 5 turns enamelled Cu wire on R6 1.4 mm L10, L11 grade 3B Ferroxcube wideband RF choke 4330 030 36642 T1 semi-rigid coax (note 4) 50 Ω length 54 mm T2, T3 semi-rigid coax (note 4) 10 Ω length 44 mm T4, T5 semi-rigid coax 25 Ω length 53 mm T6 semi-rigid coax 50 Ω length 74 mm R1 0.4 W metal film resistor 19.6 kΩ 2322 151 11963 R2 10 turn potentiometer 5 kΩ 2122 362 00725 R3, R4, R5 0.4 W metal film resistor 2.05 kΩ 2322 151 12052 R6, R7, R8 1.0 W metal film resistor 10 Ω 2322 153 71009 Notes 1. American Technical Ceramics type 100B or capacitor of same quality. 2. American Technical Ceramics type 175B or capacitor of same quality. 3. The striplines are on a double copper-clad PCB with P.T.F.E. fibre-glass dielectric (εr = 2.2); thickness 1/32 inch. 4. T2 and T3 are equipped with a Toroidal core, grade 4C6 (cat.no. 4322 020 97171). 1998 Mar 23 11, handbook, full pagewidth bias voltage RF input RF output drain voltage MGH789 Fig.6 Lay-out of one unit. 1998 Mar 23 12, handbook, full pagewidth C1 C2 R1 C3 C4 R2 MGH790 C1 = C3 = 39 pF C2 = C4 = 68 pF R1 = R2 = 5.3ΩL= 12 bonding wires per side (50 µm), heigth 450 µm Fig.7 Lay-out of BLF548 dummyload. By means of a RF-analyzer the predicted frequency response of the network can be reproduced in practice, while tuning C10 and C14 for optimum R1. This is presented in Fig.8. A comparison with the simulated networkresponse (Fig.3) shows a high amount of common behaviour. MGH791 handbook, halfpage Rl (dB) −4 −8 −12 −16 −20 100 200 300 400 500 frequency (MHz) Fig.8 Measured network response of one unit after tuning C10 and C14. 1998 Mar 23 13, 5.2 Testing the unit under RF conditions After exchanging the dummyload for a BLF548, a frequencysweep under power conditions can be made with help of a network analyzer. The used measurement set-up is given in Fig.9. handbook, full pagewidth 50ΩBALOAD

R NETWORK

RF OUTPUTANALYZER POWER BIDIRECTIONAL BIDIRECTIONAL BROADBAND BIDIRECTIONAL POWER AMPLIFIER COUPLER COUPLER AMPLIFIER COUPLER METER MGH792

COAXIAL SWITCH

INPUT POWER SPECTRUM METER ANALYZER Fig.9 Measurement set-up for frequency sweep under powerconditions. 1998 Mar 23 14, 5.3 Tuning the unit’s inputnetwork After supplying both the bias- and drain voltages to the unit and adjusting the drain quiescent current with R2 to 320 mA (160 mA per side), the inputpower (Ps) is applied. Gainflatness is optimized while tuning C7. The unit’s input returnloss is given in Fig.10. Pl versus frequency is shown in Fig.11. A comparison with the simulated network response of the inputside (Fig.4) shows a high similarity. Gainflatness within 1 dB is achieved between 100 and 450 MHz. CH1 S11 log MAG 10 dB MGH793 handbook, full pagewidth Cor Ref 0 dB (1) (1) 400 MHz; −8.3 dB. Fig.10 Input return loss of one unit (Frequency sweep at Pl = 150 W, for f = 90 to 500 MHz. 1998 Mar 23 15, CH2 S21 log MAG 1 dB MGH794 handbook, full pagBeLwFid t5h48 (1) Cor Ref 5 dB START 90 . 000 000 MHz STOP 500 . 000 000 MHz (1) 400 MHz; 12.526 dB. Fig.11 Powergain of one unit (Frequency sweep at Pl = 150 W, for f = 90 to 500 MHz). 1998 Mar 23 16, 5.4 Combining the units After tuning the second unit similar as described above, the connection was made to the 90° hybrid couplers. The couplers do contain 4 ports, one at which the input signal is applied (1). The input is devided equally into two ports (3 and 4). Between ports 3 and 4 there is a voltage lag of 90°. Mismatch at ports 3 and 4 do not effect the VSWR of port 1, since port 2 is terminated with a 50 Ω load (KDI-PPT820-75-3 flange mounted). At the output side of the amplifier the units are combined in a similar way. Both input and output hybrids and 50 Ω loads are mounted in a Brass baseplate (dimensions; 200 × 160 × 10 mm), which also serves as a heatspreader for both BLF548 devices. The baseplate is connected to a heatsink which is cooled by means of forced air. Final results are given in Fig.12. (input return loss of the amplifier) and Fig.13 (the amplifier’s powergain). CH1 S11 log MAG 5 dB MGH795 handbook, full pagewidth (1) Ref −45 dB (1) 450 MHz; −28.1 dB. Fig.12 Input return loss of complete amplifier (Frequency sweep at Pl = 250 W, for f = 90 to 500 MHz). 1998 Mar 23 17, CH2 S21 log MAG 2 dB MGH796 handbook, full pagewidth (1) Ref 0 dB START 90 . 000 000 MHz STOP 500 . 000 000 MHz (1) 450 MHz; 11.1 dB. Fig.13 Powergain of complete amplifier (Frequency sweep at Pl = 250 W, for f = 90 to 500 MHz). 6 CONCLUSIONS The described procedures shown in this paper, are a great help in designing high-power broadband amplifiers. The differences between theory and practice are relatively small. The BLF548 is very well suited to perform in multi-octave broadband UHF-amplifiers; at a supply voltage of 28 V, between 100 and 450 MHz, 250 W of outputpower could be generated with a powergain of 11 dB (gainripple smaller than 1 dB). Drainefficiency is 45 to 55% throughout the band. The reduction of the second harmonic is more than 25 dB, with respect to the fundamental. The input returnloss is better than −12 dB. 7 REFERENCES • Data Handbook SC08b, RF power MOS transistors − Philips Components • Application Report Bipolar & MOS transmitting transistors − Philips Components • A look inside those integrated two-chip amps − Joe Johnson − Microwaves feb. 1980 • Apply wideband techniques to balanced amplifiers − Lee B. Max − Microwaves apr. 1980 • Demystifying new generation silicon high power FETs − Steve McIntyre − Microwave Journal apr. 1984 • Anaren − Microwave components catalog no.17A. 1998 Mar 23 18, 8 APPENDIX A MLIN 10 20 w^w1; 1^122 DIM MBEND3 19 29 w^w1 FREQ MHz MBEND3 20 30 w^w1 RES OH MLIN 29 0 w^w1; IND NH 1^133 CAP PF MLIN 30 0 w^w1; LNG MM 1^133 VAR MCLIN68910 W = 8; !L = 13 S = 2.5; C1 = 120 !c = 270 L = 12 L2 = 74 !50 SLC 9 10 L = 0.5; !C = 5 L3 = 53 !50 C = 3 L10 = 9 !9 DEF3P1910 TRAFO !OUTPUT L11 = 0.350559 !0.5 NETWORK R11 = 4.191123 !5.1 BAL220C11 = 64.2821 !75 TRAFO31204DEF2P13IMP2 L12 = 0.409981 !0.31 BAL110W1 = 6 BAL120L22 = 10 TRAFO3120L33 = 66 DEFIP 3 IMP CKT FREQ RES10R= 50 SWEEP 50 550 25 DEFIP 1 REFIMP OUT S1PA20BLF548OU IMP RE(Z1) DEF1P 2 BALI IMP IM(Z1) IND12L^L11 IMP DB(S11) GR1 RES20R^R11 IMP S11 SC2 SLC20L^L12; !to determine zload !IMP VSWR1 C^C11 BAL3 RE(Z1) !TO DEFIP 1 BAL3 DETERMIN SLC10L= 0.5EC= 4 !13.5 BAL3 IM(Z1) !ZLOAD COAX1203DI = 0.91; TAND = 0.0002; IMP2 DB(S12) DO = 2.98; RHO = 1 L^L3; GRID ER = 2.03 RANGE 50 550 50 MSUB ER = 2.2; T = 0.035 RHO = 0.72; GR1 −3005H= 0.79 RGH = 1 TERM SLC24L= 0.5; C^cl IMP2 BAL1 !50 Ω SLC35L= 0.5; C^cl REFIMP PORT1 COAX4608DI = 1.63 DO = 2.95; L^L2 ER = 2.03; OPT TAND = 0.0002; RANGE 50 550 RHO = 1 !IMP VSWR1 < COAX5806DI = 1.63 DO = 2.95; L^L2; 1.6 ER = 2.03; IMP MODEL TAND = 0.0002; REFIMP RHO = 1 SLC68L= 0.5; !C = 41 Note C = 41 1. Slp file BLF5480U does contain data given in SLC68L= 0.7; !C = 0 “Appendix C”. C = 8 MLIN 919 w^w1; 1^122 1998 Mar 23 19, 9 APPENDIX B

Input BLF548 application (300 W/28 V/500 MHz); input in 1P file fit between 100 − 500 MHz with 1 : 4 transformer

(Zc = 10 Ω) to determine Gp and inputVSWR ZiN and Gp data derived from calculations, which represent the performance of the device after applying the output NEtwork, given in “Appendix A” to IT.

Table 5 COAX5876DI = 1.15; RHO = 1

DO = 1.45; L^L2; DIM ER = 2.03; FREQ MHz TAND = 0.0002 RES OH UNIT70IND NH SLC68L= 0.5; C\0.103064 CAP PF IC − 2.1 LNG MM MCLIN68910 W = 8; VAR S = 2.5; L\3.05628 C1\57.83880 !C = 27 SLC 9 10 L = 0.5; L2\64.28568 !L = 25 C\0.923788; L3 = 55 !L = 25 !c − 3.5 R11 = 0.00020 !4.1 DEF3P1910 TRAFO INPUT 7 NETWORK C11 = 241.805 !65 BALI209TRAFO3540L12 = 0.00004 !0.41 GPAF42GPAF51CKT DEF2P13IMP2 !TO RES10R= 50 DETERMINE DEF1P 1 REFIMP !50 Ω LOAD S11, S12 S1PA20BLF54812 FREQ DEF1P 2 BAL1 !BLF548’s Zin SWEEP 100 500 25 RES12R^R11 OUT SLC20L^L12; C^C11 IMP2 TE(Z2) DEF1P 1 BAL3 IMP2 IM(Z2) GAIN12A= 22.65; !Gp DATA IMP2 DB(S22) GR1 !S11 AT 50ΩS= −4.79; F = 150 DERIVED PORT

FROM

IMP2 DB(S12) GR1 !CALCULATED GAIN23A= 0; S = 0.5; !Table 3 POWERGAIN F = 350 IMP2 VSWR2 GAIN34A= 0; S = 1.5; !13 dB F = 400 GAINCORRE GPAF DB(S21) !BLF548’s GP CTION SINCE GRID DEF2P13GPAF !2 SECTIONS RANGE 100 500 25

ARE

INVOLVED GRI −20 20 5 SLC10L= 0.5; TERM C\0.016169 !2 IMP2 BAL1 COAX1203DI = 0.91; L^L3; REFIMP DO = 2.98 ER = 2.03; TAND = 0.000 OPT 2; RHO = 1 RANGE 150 550 MSUB ER = 2.2; IMP2 H = 0.79; DB(S12) > 11 T = 0.035; RHO = 0.72; IMP2 RHG = 1 DB(S12) < 12.4 SLC24L= 0.5 C^cl Note SLC35L= 0.5 C^cl 1. S1p file BLF548I2 does contain data given in Table 4. COAX4678DI = 1.15; RHO = 1 DO = 1.45; L^L2; Format as shown in “Appendix C”. ER = 2.03; TAND = 0.0002 1998 Mar 23 20, 10 APPENDIX C Fit dummyloadmodel to calculated BLF548 Zoutput !FIlename: E:\users\blf548ou.s1p !reference: calculated data derived from transmod program;!Zload converted to Zoutput MOD IM(Z1) DIM MES RE(Z1) FREQ GHZ MES IM(Z1) RES OH OPT IND NH MOD MES CAP PF MODEL LNG MM Table 6

CKT

# GHz MSUB ER = 6.5 H = 1.02; RI R 1Z T = 0.035 ; 0.050 5.26 −0.86 ! freq R1 X1 RHO = 1; 0.075 5.04 −1.22 RGH = 0 0.100 4.73 −1.51 IND10L= 0.168 !inductan 0.125 4.39 −1.73 ce to 0.150 4.04 −1.87 ground 0.175 3.70 −1.94 RES12R= 5.28 0 0.200 3.36 −1.95 SLC12L= 0.3 C = 40 0.225 3.05 −1.92 SLC12L= 0.3 C\65.8 0.250 2.76 −1.85 CAP20C= 1 0.275 2.50 −1.75 IND23L= 0.165 !bonding 0.300 2.27 −1.63 wires at 0.325 2.06 −1.50 drainside 0.350 1.87 −1.35 MLIN34W= 10.8 L = 1.6 !metalliza 0.375 1.71 −1.21 tion of 0.400 1.56 −1.05 header 0.425 1.43 −0.90 DEF1p 4 MOD !1-port of dummylo 0.450 1.31 −0.74 ad 0.475 1.21 −0.59 S1PA10BLF5480 !calculate 0.500 1.11 −0.43 U.s1p d 0.525 1.03 −0.28 z-parame 0.550 0.95 −0.13 ters of DEF1p 1 MES !BLF548 Note outputim 1. Zload is derived from data given in Fig.3. pedances

FREQ

SWEEP 0.1 0.5 .050

OUT

MOD RE(Z1) 1998 Mar 23 21,

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Lukiska 10, PL 04-123 WARSZAWA, Bulgaria: Philips Bulgaria Ltd., Energoproject, 15th floor, Tel. +48 22 612 2831, Fax. +48 22 612 2327 51 James Bourchier Blvd., 1407 SOFIA, Tel. +359 2 689 211, Fax. +359 2 689 102 Portugal: see Spain Canada: PHILIPS SEMICONDUCTORS/COMPONENTS, Romania: see Italy Tel. +1 800 234 7381 Russia: Philips Russia, Ul. Usatcheva 35A, 119048 MOSCOW, China/Hong Kong: 501 Hong Kong Industrial Technology Centre, Tel. +7 095 755 6918, Fax. +7 095 755 6919 72 Tat Chee Avenue, Kowloon Tong, HONG KONG, Singapore: Lorong 1, Toa Payoh, SINGAPORE 1231, Tel. +852 2319 7888, Fax. +852 2319 7700 Tel. +65 350 2538, Fax. +65 251 6500 Colombia: see South America Slovakia: see Austria Czech Republic: see Austria Slovenia: see Italy Denmark: Prags Boulevard 80, PB 1919, DK-2300 COPENHAGEN S, South Africa: S.A. PHILIPS Pty Ltd., 195-215 Main Road Martindale, Tel. +45 32 88 2636, Fax. +45 31 57 0044 2092 JOHANNESBURG, P.O. Box 7430 Johannesburg 2000, Finland: Sinikalliontie 3, FIN-02630 ESPOO, Tel. +27 11 470 5911, Fax. +27 11 470 5494 Tel. +358 9 615800, Fax. +358 9 61580920 South America: Al. Vicente Pinzon, 173, 6th floor, France: 51 Rue Carnot, BP317, 92156 SURESNES Cedex, 04547-130 SÃO PAULO, SP, Brazil, Tel. +33 1 40 99 6161, Fax. +33 1 40 99 6427 Tel. +55 11 821 2333, Fax. +55 11 821 2382 Germany: Hammerbrookstraße 69, D-20097 HAMBURG, Spain: Balmes 22, 08007 BARCELONA, Tel. +49 40 23 53 60, Fax. +49 40 23 536 300 Tel. +34 3 301 6312, Fax. +34 3 301 4107 Greece: No. 15, 25th March Street, GR 17778 TAVROS/ATHENS, Sweden: Kottbygatan 7, Akalla, S-16485 STOCKHOLM, Tel. +30 1 4894 339/239, Fax. +30 1 4814 240 Tel. +46 8 632 2000, Fax. +46 8 632 2745 Hungary: see Austria Switzerland: Allmendstrasse 140, CH-8027 ZÜRICH, Tel. +41 1 488 2686, Fax. +41 1 488 3263 India: Philips INDIA Ltd, Band Box Building, 2nd floor, 254-D, Dr. Annie Besant Road, Worli, MUMBAI 400 025, Taiwan: Philips Semiconductors, 6F, No. 96, Chien Kuo N. Rd., Sec. 1, Tel. +91 22 493 8541, Fax. +91 22 493 0966 TAIPEI, Taiwan Tel. +886 2 2134 2865, Fax. +886 2 2134 2874 Indonesia: see Singapore Thailand: PHILIPS ELECTRONICS (THAILAND) Ltd., 209/2 Sanpavuth-Bangna Road Prakanong, BANGKOK 10260, Ireland: Newstead, Clonskeagh, DUBLIN 14, Tel. +66 2 745 4090, Fax. +66 2 398 0793 Tel. +353 1 7640 000, Fax. +353 1 7640 200 Turkey: Talatpasa Cad. No. 5, 80640 GÜLTEPE/ISTANBUL, Israel: RAPAC Electronics, 7 Kehilat Saloniki St, PO Box 18053, Tel. +90 212 279 2770, Fax. +90 212 282 6707 TEL AVIV 61180, Tel. +972 3 645 0444, Fax. +972 3 649 1007 Ukraine: PHILIPS UKRAINE, 4 Patrice Lumumba str., Building B, Floor 7, Italy: PHILIPS SEMICONDUCTORS, Piazza IV Novembre 3, 252042 KIEV, Tel. +380 44 264 2776, Fax. +380 44 268 0461 20124 MILANO, Tel. +39 2 6752 2531, Fax. +39 2 6752 2557 United Kingdom: Philips Semiconductors Ltd., 276 Bath Road, Hayes, Japan: Philips Bldg 13-37, Kohnan 2-chome, Minato-ku, TOKYO 108, MIDDLESEX UB3 5BX, Tel. +44 181 730 5000, Fax. +44 181 754 8421 Tel. +81 3 3740 5130, Fax. +81 3 3740 5077 United States: 811 East Arques Avenue, SUNNYVALE, CA 94088-3409, Korea: Philips House, 260-199 Itaewon-dong, Yongsan-ku, SEOUL, Tel. +1 800 234 7381 Tel. +82 2 709 1412, Fax. +82 2 709 1415 Uruguay: see South America Malaysia: No. 76 Jalan Universiti, 46200 PETALING JAYA, SELANGOR, Tel. +60 3 750 5214, Fax. +60 3 757 4880 Vietnam: see Singapore Mexico: 5900 Gateway East, Suite 200, EL PASO, TEXAS 79905, Yugoslavia: PHILIPS, Trg N. Pasica 5/v, 11000 BEOGRAD, Tel. +9-5 800 234 7381 Tel. +381 11 625 344, Fax.+381 11 635 777 Middle East: see Italy For all other countries apply to: Philips Semiconductors, Internet: http://www.semiconductors.philips.com International Marketing & Sales Communications, Building BE-p, P.O. Box 218, 5600 MD EINDHOVEN, The Netherlands, Fax. +31 40 27 24825 © Philips Electronics N.V. 1998 SCA57 All rights are reserved. Reproduction in whole or in part is prohibited without the prior written consent of the copyright owner. The information presented in this document does not form part of any quotation or contract, is believed to be accurate and reliable and may be changed without notice. No liability will be accepted by the publisher for any consequence of its use. Publication thereof does not convey nor imply any license under patent- or other industrial or intellectual property rights. Printed in The Netherlands Date of release: 1998 Mar 23]

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