Download: APPLICATION NOTE A wide-band linear power amplifier (470 − 860 MHz) with two transistors BLW34 ECO7901

APPLICATION NOTE A wide-band linear power amplifier (470 − 860 MHz) with two transistors BLW34 ECO7901,

CONTENTS

1 ABSTRACT 2 INTRODUCTION 3 THEORETICAL CONSIDERATIONS 3.1 The equivalent circuit of the BLW34 3.1.1 The output network 3.1.2 The input network 4 THE HYBRID COUPLED AMPLIFIER 4.1 Practical considerations 4.2 Practical optimization 4.3 Intermodulation, VSWR and gain measurements 5 CONCLUSIONS 6 REFERENCES 1998 Mar 23 2, 1 ABSTRACT For application in driver or final stages of TV-transposers in band IV/V (470-860 MHz) a linear wideband power amplifier has been designed with 2 transistors BLW34 coupled by means of 3 dB −90° hybrids. Each transistor is adjusted in class-A at VCE = 25 V and Ic = 0.6 A. The peak sync output power for a 3-tone I.M. distortion of −60dB varies between 3.6 and 5.4 W. The power gain is 9.1 ± 0.3 dB. Input and output VSWR are below 1.3. 2 INTRODUCTION This report describes the realisation of a wide-band UHF power amplifier for TV transposer service in band IV and V (470 − 860 MHz). The amplifier is designed with the BLW34 transistor being developed for ultra linear applications operating in class A. Each device is able to deliver at least 1.8 W peak sync output. The BLW34 forms a series with the smaller devices BLW33 (1.0 W) and BLW32 (0.5 W). The power gain at 860 MHz is at least 9 dB. The BLW34 is encapsulated in a 1/4 inch capstan envelope with ceramic cap. 3 THEORETICAL CONSIDERATIONS 3.1 The equivalent circuit of the BLW34 For class A operation the BLW34 is specified at VCE = 25 V; Ic = 600 mA. The corresponding typical gain, input and load impedance according to the Data sheets are given in Table 1. Table1fGAIN Ri (SERIES) Xi (SERIES) RL (SERIES) XL (SERIES) (MHz) (dB) (Ω) (Ω) (Ω) (Ω) 470 15.2 1.46 1.93 12.8 11.0 636 12.6 1.39 2.84 8.85 9.97 860 10.1 1.27 4.00 5.36 7.67 To facilitate calculations an approximate equivalent circuit for the transistor input and output impedance can be given. It is shown in Fig.1. handbook, h0a.lf7p1a gneH 0.57 nHBCic 1.4 Ω 26 Ω 13.5 pFEEMGL353 Fig.1 1998 Mar 23 3, 3.1.1 THE OUTPUT NETWORK The circuit will be designed on printed circuit board with PTFE fibre glass as a dielectric having an εr = 2.74 and thickness of 1/32 inch. The input and output network start with a piece of stripline having a width of 6 mm, being the width of the base and collector leads. For a dielectric of 1/32 inch the characteristic resistance is 21 Ω. The length for the collector lead amounts to 3 mm, but the base lead is different in length. The first step in the matching is to tune out the output capacitance of the transistor by means of the collector R.F. choke. This choke is executed as a stripline with a width of 1 mm corresponding with a characteristic resistance of 72 Ω to keep the parallel capacitance at this point as low as possible. For practical reasons the choke is connected to the main transmission line at a distance of 3 mm from the transistor edge. For the design procedure one is referred to Part 1 of this handbook (SC19B). The results of the calculations before and after computer optimization are given in Fig.2 and Table 2. 21.2 Ω 71.7 Ω 71.7 Ω handbook, full pagewidth 0.57 nH4.67 mm l2 C l3 1 C2 ZL l1 50 Ω 0.5 nH 0.9 nH 26 Ω 13.5 pF MGL352 Fig.2 Table 2 ELEMENT BEFORE OPTIM. AFTER OPTIM. UNIT l1 16.8 22.1 mm l2 18.2 20.8 mm C1 8.58 9.76 pF l3 38.2 43.5 mm C2 3.91 3.09 pF Smax. 2.48 1.23 − Smax is the maximum VSWR of the network. The lenghts given hold for air lines. The actual lengths on the printed-circuit board are shorter; the reduction factors are: 1.445 for the 72 Ω lines and 1.556 for the 21 Ω line. 1998 Mar 23 4, The predicted minimum output power of the complete amplifier with two transistors BLW34 is: 2Po Po 1 ⋅ 0.95 2 ⋅ 1.82 = - = - ⋅ 0.95 ≅ 2.8WSmax 1.23 Po1 is the minimum output power of one transistor in a narrow band circuit, Smax is as specified above and the factor 0.95 represents the power loss of the hybrid coupler at the output. 3.1.2 THE INPUT NETWORK The design procedure can be found in Part 1 of this handbook (SC19B). The results of the calculations are summarized in Fig.3 and Table 3. handbook, full pagewidth 71.7 Ω 21.2 Ω l1C1 l2 C2 0.71 nH 50 Ω 0.9 nH 0.5 nH 1.4 Ω MGL354 Fig.3 Table 3 ELEMENT BEFORE OPTIM. AFTER OPTIM. UNIT C1 4.78 4.41 pF l1 25.9 32.1 mm C2 21.9 19.9 pF l2 13.3 10.5 mm ∆G ±1.93 ±0.12 dB ∆G is the resulting power gain variation caused by the transistor and the input network over the frequency band. The lengths of the lines hold for air as a dielectric. Transformation to striplines on a printed-circuit board is done in the same way as in the previous section. The minimum power gain of the complete amplifier with two transistors BLW34 is expected to be: Go = Gt − 2AH − A1 − A2 1998 Mar 23 5, in which: Gt = minimum power gain of BLW34 in a narrow band circuit. AH = power loss of one hybrid coupler A1 = reflection loss of input network A2 = reflection loss of output network. In practical figures this means: Go = 9.0 − 2 × 0.2 − 0.15 − 0.05 = 8.4 dB. The input VSWR of a single amplifier was calculated to vary from 11 at 470 MHz down to 1.45 at 860 MHz. 4 THE HYBRID COUPLED AMPLIFIER 4.1 Practical considerations On previous pages the theoretical approach of a single amplifier has been discussed. In practice, it was the intention to realize a small compact amplifier on a printed-circuit board with the input and output terminal (Rc = 50 Ω) in line for easy cascading of several amplifiers. Besides the wide-band properties it is the intention to obtain a higher output power, so two BLW34 branches are connected in parallel. At the same time it is of course rather unacceptable that the amplifier loads a driver stage with a mismatch causing a VSWR of 11 at 470 MHz. Both problems may be solved sufficiently by applying two BLW34 branches in parallel with the aid of two wide-band 3 dB −90 °C. coaxial hybrids on a 50 Ω basis. In this configuration the output power will be nearly doubled; reduction of the input VSWR of the complete system to a value of around 1.2 may be explained from the properties of the coupler applied. The reflected power will be absorbed in the resistor matching the isolated port. This resistor is 50 Ω and consists of two 100 Ω power metal film resistor in parallel. The same has been done on the output side. The printed-circuit board needs to be double copper clad with a PTFE fibre glass dielectric (εr = 2.74) for low losses at UHF. A thickness of 1/32 inch has been chosen; so the 72 Ω lines are 1 mm wide. Figure 4 shows the circuit diagram of the complete 2× BLW34 class A amplifier. The biasing circuit is drawn in Fig.5. The printed-circuit board and the amplifier lay-out in Fig.6. For a correct earthing the upper earth sheet parts are connected to the lower sheet by soldering copper straps at the edges of the printed-circuit board. The black parts in Fig.6 are the soldered copper straps. The emitters are grounded as short as possible by applying copper straps under the emitter leads. For that reason the holes in the board are square instead of round. Both transistors are screwed to a water-cooled brass heatsink. So, several heatsink temperatures can be applied by means of a TAMSON unit supplying water with a thermostatically controlled temperature. The tuning capacitors in the circuit are of the film dielectric type with three tags of which both earth terminals are fed through small holes and soldered to the upper as well as the lower plane. Fixed capacitors in the r.f. path are of the multi-layer ceramic chip type. The coaxial connectors are of the SMA 50 Ω type, being soldered to upper and lower sheet. 4.2 Practical optimization We started with optimization on a small signal basis with the circuit inserted in a network analyzer chain, having swept S-parameter facilities. 1998 Mar 23 6, So far, similar tuning methods have been applied as described in Ref. 2. Because it is rather complicated to find the best compromise between an acceptable flat gain curve (S21) and sufficient output power with low i.m.d. a dynamic large signal optimization method has been realized. Figure 7 shows the block diagram. This tuning method is based on the correlation between the single tone 1dB compression point and the i.m.d figure of a linear amplifier. In this set-up the swept output power level of the amplifier under test is kept constant and the required (detected) drive power monitored on an oscilloscope screen (PM3260). The swept drive power is available from the sweep generator HP8620C in combination with RF plug-in unit HP86222A. Because the output power of this system is too low viz. appr. 20 mW (+13 dBm) a combined amplifier with BLW32 and BLW33 (Ref. 2) has been added. The latter combination shows an overall gain of approx. 20 db in the range 470 − 860 MHz. When the circuit under test is inserted in the chain of Fig.7, the input power measured on port C of hybrid 1 corresponds in principle with the gain curve (S21) being measured with the network analyzer; in fact one is the inverse of the other. When the drive level is slowly increased, the shape of the gain curve changes somewhat when compression starts. By careful retuning of the amplifier the shape of the gain curve can be corrected again in the direction of the original smaller signal curve. The actual single tone output power has been measured with the aid of the calorimetric watt meter HP435A when the action of the sweep is stopped. Also it is possible to examine the output signal itself by means of a spectrum analyzer being loosely coupled via a 50 Ω pick-off device. Resuming it can be said that the advantage of applying this high level tuning system is characterized by the fact that the output power is leveled and so compression does not start earlier due to gain fluctuations of the amplifier. This makes the judgement of the compression level easier. 4.3 Intermodulation, VSWR and gain measurements For i.m.d. measurements on television systems the post offices advice and apply the 3-tone test method (vision carrier −8 dB, sound carrier −7 dB, sideband signal −16 dB; zero dB corresponds to peak sync level). For this reason a wideband test set-up has been realized. The block diagram is in Fig.8. In this set-up first the sound and vision carriers are joined in the wide-band coaxial hybrid H1.Then the sideband signal from the (smaller) generator SMLU is added in hybrid H2. The complete 3-tone signal passes a low-pass filter (700 or 1000 MHz cut-off depending on input frequency) a continuously variable attenuator and a circulator (three different types needed to cover at least the range 470 − 860 MHz). The output power is measured with a HP435A calorimetric watt meter and the i.m.d. observed with a spectrum analyzer (HP8558B). Figure 9 shows the 3-tone i.m.d. results measured on the complete hybrid coupled 2 × BLW34 amplifier. To get an idea of the Po sync drop for different heatsink temperatures, measurements have been done for THS = 23 °C and THS = 70 °C. The maximum drop in Po sync amounts to 1.1 dB. Besides the 3-tone test the peak envelope power has been measured in a two-tone way for a third order i.m.d of d3 = −47 dB. It can be proved that there is a correlation between the afore described −60 dB 3-tone and the −47 dB two-tone test method. It is known that the advantage of video precorrection on i.m.d. amounts to values up to 10 dB. Calculating with an average of 8 dB, the two-tone i.m.d. for d3 = 39 dB is interesting. From Fig.10 it may be seen that the output power is more than doubled. Both tests have been done for heatsink temperatures of resp. 23 and 70 °C. 1998 Mar 23 7, Finally input and output VSWR and gain figures were measured only under small signal conditions. The results of amplifier branches 1 and 2 for a heatsink temperature of 70 °C are shown in Figs 11 to 13. The VSWR figures of the input and output are expressed in reflection damping (resp. S11 and S22) on a 50 Ω basis. According to Fig.11 the minimum reflection damping of the input amounts to S11 = −1 dB (VSWR = 17.4) at 470 MHz and to S11 = −10 dB (VSWR = 1.93) at 860 MHZ. On the output side the worst reflection damping amounts to S22 = −5.5 dB (VSWR = 3.26) in the middle of the passband (Fig.13). The power gain, represented by S21, is appr. 9.5 dB (Fig.12). Resuming it can be said that, as Figs 11 to 13 show, the differences in gain, input and output VSWR between both branches are rather small. Figs 14 to 16 show the final results of the complete hybrid coupled amplifier. Examining the S11 and S22 figures it appears that the minimum input reflection damping (S11) amounts to appr. 18 dB, what corresponds with a maximum VSWR of 1.29. The latter value may be mainly explained from the specified maximum VSWR = 1.25 of the applied Anaren hybrids. If more attention is paid to matching of the isolated ports the results may be improved. The matching consists of two metal film power transistors of 100 Ω in parallel. They have low-frequency tolerances of 5%. The small signal gain of the complete amplifier amounts to 9.1 ±0.3 dB for THS = 23 °C, whilst the dashed line shows the results for THS = 70 °C being 8.9 ± 0.3 dB. The temperature influence on the S11 and S22 figures is almost negligible. 5 CONCLUSIONS On preceding pages the theoretical and practical design has been described of the wide-band (470 − 860 MHz) high quality linear amplifier being equipped with two BLW34 transistors operating in class A. There are some small differences between the theoretical design and the practical circuit. • The calculated value for the chip capacitors C11 = C12 = C13 = C14 in Fig.4 was 10 pF. In practice 8.2 pF appeared to be a better choice. • Also the values of C19 = C20 = C21 = C22, being calculated as 4.7 pF are changed. The new value is 3.9 pF. • In the first instance a chip capacitor of 1.5 pF in parallel with C23 and C26 was planned. However this capacitor could be omitted in the operational circuit. The excepted results for a single amplifier: Gp min. = 8.8 dB; and Po sync = 1.5 W at a 3-tone i.m.d. of −60 dB and Gp min = 8.4 dB and Po sync = 2.8 W for THS ≤ 70 °C are realized in this design. Here, we have calculated with a total insertion loss of 0.4 dB for the applied hybrids. They are specified for a maximum of 0.25 dB per device (1.06 × power). 6 REFERENCES Ref. 1: O. Pitzalis Jr. and R.A. Gilson - Tables of impedance matching networks which approximate prescribed attenuation versus frequency slopes. IEEE Transactions on microwave theory and techniques, Vol. MTT-19, no. 4, April 1971, pp. 381-386. Ref. 2: A.H. Hilbers and M.J. Köppen - Wide-band linear power amplifier (470 − 860 MHz) with the transistors BLW32 and BLW33. C.A.B. report ECO7806. 1998 Mar 23 8, handbook, full pagewidth Vb1 C7 R3 C9 C15 C17 L3 +VC1 L9 C3 C11 BLW34 C19 C23 C1L1 L11 L13 L4 L7 R5 R6 C4 C12 C20 C24 50 Ω in 0° −90° iso input H1 H2 iso −90° 0° in 50 Ω output C5 C13 C21 C25 R1 R2 C2L5 L8L2 L12 L14 C6 C14 BLW34 C22 C26 L10 MGL355 L6 +VC2 C8 R4 C10 C16 C18 Vb2 Fig.4 1998 Mar 23 9, handbook, full pagewidth BLW34 +VC R10 +VS R8 C27 +V C29B R11 R12 C28 R13 BAW62 R9 BD138 R14 R7 MGL357 Fig.5 1998 Mar 23 10, Table 4 LIST OF COMPONENTS C1 = C2 = 2.2 pF multilayer ceramic chip capacitor Tekelec-Airtronic part no. 500 R15 N2R2 BA C3 = C6 = C23 = C26 = 1.4 to 5.5 pF film dielectric trimmer (cat. no. 2222 809 09001) C4 = C5 = C24 = C25 = 100 pF multilayer ceramic chip capacitor (cat. no. 2222 851 13101) C7 = C8 = C29 = 100 nF polyester capacitor C9 = C10 = C15 = C16 = 100 pF multilayer ceramic chip capacitor (cat. no. 2222 852 13101) C11 = C12 = C13 = C14 = 8.2 pF multilayer ceramic chip capacitor, ATC (American Technical Ceramics) type 100A-8R2-J-Px-50 C17 = C18 = C28 = 6.8 µF, 63 V electrolytic capacitor C19 = C20 = C21 = C22 = 3.9 pF multilayer ceramic chip capacitor, Tekelec-Airtronic part no. 500 R15 N3R9 CA C27 = 470 nF polyester capacitor R1 = R2 = R5 = R6 = 100 Ω (±5%) power metal film resistor PR37 type (cat. no. 2322 191 31001) R3 = R4 = R8 = 10 Ω (±5%) carbon resistor; CR25 type R7 = 33 Ω (±5%) carbon resistor; CR25 type R9 = 220 Ω (±5%) power metal film resistor PR52 type (cat. no. 2322 192 32201) R10 = 5.6 Ω (±5%) enamelled wire-wound resistor WR0617E style R11 = 8.2 Ω (±5%) enamelled wire-wound resistor WR0617E style R12 = 100 Ω cermet preset potentiometer R13 = 120 Ω (±5%) carbon resistor; CR25 type R14 = 1.8 kΩ (±5%) carbon resistor; CR25 type H1 = H2 = ultra-miniature 3 dB −90° coupler model no. 10264-3, range 0.5 − 1,0 GHz; Anaren Microwave Inc. L1 = L2 stripline (Zc = 72 Ω), 22.1 × 1.0 mm2; note 1 L3 = L6 = 1 µH microchoke L4 = L5 stripline (Zc = 21 Ω), 6.7 × 6.0 mm2; note 1 L7 = L8 = stripline (Zc = 21 Ω), 3.0 × 6.0 mm2; note 1 L9 = L10 = stripline (Zc = 72 Ω), 15.2 × 1.0 mm2; note 1 L11 = L12 = stripline (Zc = 72 Ω), 14.3 × 1.0 mm2; note 1 L13 = L14 = stripline (Zc = 72 Ω), 29.9 × 1.0 mm2; note 1 Note 1. These striplines are printed on double Cu-clad printed-circuit board with PTFE fibre-glass dielectric (εr = 2.74); thickness 1/32”. 1998 Mar 23 11, handbook, full pagewidth 107 50 Ω R6 R5 L13 L14 C17 C19 C21 C18 C20 C22 C15 C24 C25 C16 L11 L12 L9 C23 C26 L10C C C9 C3 C6 C10 L5 L4 L6 R3 C11 C12 C4 C5 0° ° C13 C14C1 −90 C2 R4 L1 ANAREN L2 10264-3 C7 5 - 1 GHz C8 IN ISO R2 R1 MGM805 50 Ω

Fig.6 Printed-circuit board and amplifier lay-out.

1998 Mar 23 12 0° −90°

ANAREN

10264-3 5 - 1 GHz IN ISO, H−183−4 coaxial det.handbook, full pagewidth 50 Ω ATT. AACSCOPE 19 dB PM3260 H1 B HP8620C HP86222A r.f. SWEEP OSC. RF PLUG-IN BLW32 BLW3BDout sweep ext. ALC H−183−4 VAR ATT. ATT. DB6dB D.U.T.25 dB coaxial det. H22xBLW34 CAL.W. pick-off ATT. 50 ΩC A MTR. 3 dB HP435A gain curve for constant output power SPECTRUM 0−line ANAL. HP8558B 470 860 MHz MGL358 Fig.7 Dynamic compression test set-up. 1998 Mar 23 13, handbook, full pagewidth 0 dB peak sync level 50 Ω −8 dB −7 dB (vision) > 60 dB −16 dB GENERATOR −8 dB H−183−4 ampl. SLRD 50ΩCA50ΩfH1 vision i.m.d sideband sound (sound) H−183−4 carrier product signal carrier GENERATOR −7 dB 50 ΩDBCA

SLRD

50 Ω H2 50 Ω (sideband) D B LOW-PASS VAR ATT. GENERATOR −16 dB quality SMLU 700-1000 MHz AJ21ON −75 dB (470-860) CAL.W. pick-off MHzD.U.T. MTR. HP435A2xBLW34

SPECTRUM

ANAL. HP8558B MGL359 Circulator ranges: (1) 470 − 600 MHz. (2) 600 − 800 MHz. (3) 790 − 1000 MHz. Fig.8 3-tone test set-up 470 − 860 MHz. 1998 Mar 23 14, MGL367 6.0 handbook, full pagewidth Po sync2xBLW34 hybrid coupled (W) three-tone i.m.d. 5.0 THS = 23 °C −60 dB 4.0 3.0 THS = 70 °C −60 dB 2.0 1.0 450 500 550 600 650 700 750 800 850 900 f (MHz) Fig.9 MGL368 handbook, full pagewidth P PEP2xBLW34 hybrid coupledo T = 23 °C two-tone i.m.d.(W) HS 10 d3 = −39 dB 8 THS = 70 °C d3 = −39 dB THS = 23 °C 4 d3 = −47 dB THS = 70 °C 2 d3 = −47 dB 450 500 550 600 650 700 750 800 850 900 f (MHz) Fig.10 1998 Mar 23 15, MGL361 handbook, full pagewidth (dB) −2 −4 −6 −8 −10 S11 −12 450 500 550 600 650 700 750 800 850 900 f (MHz) Fig.11 MGL362 handbook, full pagewidth (dB) S21 450 500 550 600 650 700 750 800 850 900 f (MHz) THS = 70 °C. V = 25 V. Branch 1.CE IC = 600 mA. Branch 2. Fig.12 1998 Mar 23 16, MGL363 handbook, full pagewidth (dB) −2 −4 −6 −8 −10 −12 S22 −14 450 500 550 600 650 700 750 800 850 900 f (MHz) THS = 70 °C. V = 25 V. Branch 1.CE IC = 600 mA. Branch 2. Fig.13 MGL364 handbook, full pagewidth (dB) −4 −8 −12 −16 −20 −24 −28 S11 −32 450 500 550 600 650 700 750 800 850 900 f (MHz) THS = 23 °C. THS = 70 °C. Fig.14 1998 Mar 23 17, MGL365 handbook, full pagewidth (dB) S21 450 500 550 600 650 700 750 800 850 900 f (MHz) Fig.15 MGL366 handbook, full pagewidth (dB) −4 −8 −12 −16 −20 −24 −28 S22 −32 450 500 550 600 650 700 750 800 850 900 f (MHz) THS = 23 °C. THS = 70 °C. Fig.16 1998 Mar 23 18,

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